Chapter 4

Op-Amps and their Properties

Audio design has for many years relied on a very small number of op-amp types; the TL072 and the 5532 dominated the audio small-signal scene for many years. The TL072, with its JFET inputs, was used wherever its negligible input bias currents and low cost were important. For a long time the 5534/5532 was much more expensive than the TL072, so the latter was used wherever feasible in an audio system, despite its inferior noise, distortion, and load-driving capabilities. The 5534 was reserved for critical parts of the circuitry. Although it took many years, the price of the 5534 is now down to the point where you need a very good reason to choose any other type of op-amp for audio work.

The TL072 and the 5532 are dual op-amps; the single equivalents are TL071 and 5534. Dual op-amps are used almost universally, as the package containing two is usually cheaper than the package containing one, simply because it is more popular.

There are, however, other op-amps, some of which can be useful, and a selected range is covered here.

A Very Brief History of Op-Amps

The op-amp is today thought of as quintessentially a differential amplifier, responding to the difference of the input voltages while (hopefully) ignoring any common-mode component. The history of differential amplifiers goes back to that great man Alan Blumlein, and his 1936 patent [1] for a pair of valves with their cathodes connected to ground through a common resistor. However, the first valve-based operational amplifiers, i.e. those intended to be capable of performing a mathematical operation, were in fact not differential at all, having only one input. That had to be an inverting input, of course, so you could apply negative feedback.

The first op-amp to get real exposure in the UK was the Fairchild uA709, designed by the renowned Bob Widlar and introduced in 1965. It was a rather awkward item that required quite complicated external compensation and was devoid of output short-circuit protection. One slip of the probe and an expensive IC was gone. It was prone to latch-up with high common-mode voltages and did not like capacitative loads. I for one found all this most discouraging, and gave up on the 709 pretty quickly. If you’re going to quit, do it early, I say.

The arrival of the LM741 was a considerable relief. To my mind, it was the first really practical op-amp, and it was suddenly possible to build quite complex circuitry with a good chance of it being stable, doing what it should do, and not blowing up at the first shadow of an excuse. I have given some details of it in this chapter for purely historical reasons. There is also an interesting example of how to apply the LM741 appropriately in Chapter 17.

The first IC op-amps opened up a huge new area of electronic applications, but after the initial enthusiasm for anything new, the audio market greeted these devices with less than enthusiasm. There were good reasons for this. The LM741 worked reliably; the snag with using it for audio was the leisurely slew rate of 0.5 V/μs, which made full output at 20 kHz impossible. For a period of at least 5 years, roughly from 1972 to 1977, the only way to obtain good performance in a preamp was to stick with discrete transistor Class-A circuitry, and this became recognized as a mark of high quality. The advent of the TL072 and the 5532 changed this situation completely, but there is still marketing cachet to be gained from a discrete design.

An excellent and detailed history of operational amplifiers can be found in Ref. [2].

Op-Amp Properties: Noise

There is no point in regurgitating manufacturers’ data sheets, especially since they are readily available on the internet. Here I have simply ranked the op-amps most commonly used for audio in order of voltage noise (Table 4.1).

Table 4.1   Op-amps ranked by voltage noise density (typical)

The great divide is between JFET input op-amps and BJT input op-amps. The JFET op-amps have more voltage noise but less current noise than bipolar input op-amps, the TL072 being particularly noisy. If you want the lowest voltage noise, it has to be a bipolar input. The difference, however, between a modern JFET-input op-amp such as the OPA2134 and the old faithful 5532 is only 4 dB, but the JFET part is a good deal more costly. The bipolar AD797 seems to be out on its own here, but it is a specialized and expensive part. The LT1028 is not suitable for audio use for reasons described later. The LM741, which is included in this chapter for purely historical reasons, is omitted from Table 4.1 because there are no noise specs on its data sheets.

Op-amps with bias-cancellation circuitry are normally unsuitable for audio use due to the extra noise this creates. The amount depends on circuit impedances, and is not taken into account in Table 4.1. The general noise behavior of op-amps in circuits is dealt with in Chapter 1.

Op-Amp Properties: Slew Rate

Slew rates vary more than most parameters; a range of 100:1 is shown in Table 4.2. The slowest is the 741, which is the only type not fast enough to give full output over the audio band. There are faster ways to handle a signal, such as current-feedback architectures, but they usually fall down on linearity. In any case, a maximum slew rate greatly in excess of what is required appears to confer no benefits whatever.

Table 4.2   Op-amps ranked by slew rate (typical)

Op-amp

V/μs

LM741

0.5

OP270

2.4

OP27

2.8

NJM4556

3

MC33078

7

LM833

7

5532A

9

LT1028

11

TL072

13

5534A

13

OPA2134

20

LM4562

20

AD797

20

OP275

22

OPA604

25

OPA627

55

The 5532 slew rate is typically ±9 V/μs. This version is internally compensated for unity-gain stability, not least because there are no spare pins for compensation when you put two op-amps in an eight-pin dual package. The single-amp version, the 5534, can afford a couple of compensation pins, and so is made to be stable only for gains of 3× or more. The basic slew rate is therefore higher at ±13 V/μs.

Compared with power-amplifier specs, which often quote 100 V/μs or more, these speeds may appear rather sluggish. In fact they are not; even ±9 V/μs is more than fast enough. Assume you are running your op-amp from ±18 V rails, and that it can give a ±17 V swing on its output. For most op-amps this is distinctly optimistic, but never mind. To produce a full-amplitude 20 kHz sine wave you only need 2.1 V/μs, so even in the worst case there is a safety margin of at least four times. Such signals do not of course occur in actual use, as opposed to testing. More information on slew limiting is given in the section on op-amp distortion.

Op-Amp Properties: Common-Mode Range

This is simply the range over which the inputs can be expected to work as proper differential inputs. It usually covers most of the range between the rail voltages, with one notable exception. The data sheet for the TL072 shows a common-mode (CM) range that looks a bit curtailed at −12 V. This bland figure hides the deadly trap this IC contains for the unwary. Most op-amps, when they hit their CM limits, simply show some sort of clipping. The TL072, however, when it hits its negative limit, promptly inverts its phase, so your circuit either latches up, or shows nightmare clipping behavior with the output bouncing between the two supply rails. The positive CM limit is, in contrast, trouble-free. This behavior can be especially troublesome when TL072s are used in high-pass Sallen-and-Key filters.

Op-Amp Properties: Input Offset Voltage

A perfect op-amp would have its output at 0 V when the two inputs were exactly at the same voltage. Real op-amps are not perfect and a small voltage difference – usually a few millivolts – is required to zero the output. These voltages are large enough to cause switches to click and pots to rustle, and DC blocking is often required to keep them in their place. This issue is examined in depth in Chapter 11.

The typical offset voltage for the 5532A is ±0.5 mV typical, ±4 mV maximum at 25ºC; the 5534A has the same typical spec but a lower maximum at ±2 mV. The input offset voltage of the new LM4562 is only ±0.1 mV typical, ±4 mV maximum at 25ºC.

Op-Amp Properties: Bias Current

Bipolar-input op-amps not only have larger noise currents than their JFET equivalents; they also have much larger bias currents. These are the base currents taken by the input transistors. This current is much larger than the input offset current, which is the difference between the bias current for the two inputs. For example, the 5532A has a typical bias current of 200 nA, compared with a much smaller input offset current of 10 nA. The LM4562 has a lower bias current of 10 nA typical, 72 nA maximum. In the case of the 5532/4 the bias current flows into the input pins as the input transistors are NPN.

Bias currents are a considerable nuisance; when they flow through variable resistors they make them noisy when moved. They will also cause significant DC offsets when they flow through high-value resistors.

It is often recommended that the effect of bias currents can be canceled out by making the resistance seen by each op-amp input equal. Figure 4.1(a) shows a shunt-feedback stage with a 22 kΩ feedback resistor. When 200 nA flows through this it will generate a DC offset of 4.4 mV, which is rather more than we would expect from the input offset voltage error.

Figure 4.1: Compensating for bias-current errors in a shunt-feedback stage. The compensating resistor must be bypassed by a capacitor C2 to prevent it adding Johnson noise to the stage

If an extra resistance Rcompen, of the same value as the feedback resistor, is inserted into the non-inverting input circuit then the offset will be canceled. This strategy works well and is done almost automatically by many designers. However, there is a snag. The resistance Rcompen generates extra Johnson noise, and to prevent this it is necessary to shunt the resistance with a capacitor, as in Figure 4.1(b). This extra component costs money and takes up PCB space, so it is questionable if this technique is actually very useful for audio work. It is usually more economical to allow offsets to accumulate in a chain of op-amps, and then remove the DC voltage with a single output blocking capacitor. This assumes that there are no stages with a large DC gain, and that the offsets are not large enough to significantly reduce the available voltage swing. Care must also be taken if controls are involved, because even a small DC voltage across a potentiometer will cause it to become crackly, especially as it wears.

FET input op-amps have very low bias current at room temperature; however, it doubles for every 10ºC rise. This is pretty unlikely to cause trouble in most audio applications, but a combination of high internal temperatures and high-value pots could lead to some unexpected crackling noises.

Op-Amp Properties: Cost

While it may not appear on the data sheet, the price of an op-amp is obviously a major factor in deciding whether or not to use it. Table 4.3 was derived from the averaged prices for 1+ and 25+ quantities across a number of UK distributors. At the time of writing (September 2009) the cheapest popular op-amps are the TL072 and the 5532, and these happened to come out at exactly the same price, so their price is taken as unity and used as the basis for the price ratios given.

Table 4.3   Op-amps ranked by price (2009) relative to 5532 and TL072

Table 4.3 was compiled using prices for DIL packaging and the cheapest variant of each type. Price is per package and not per op-amp section. It is obviously only a rough guide. Purchasing in large quantities or in different countries may change the rankings somewhat (even going from 1+ to 25+ causes some changes) but the basic look of things will not alter too much. One thing is obvious – the 5532 is one of the great op-amp bargains of all time.

Op-Amp Properties: Distortion

Relatively few discussions of op-amp behavior deal with non-linear distortion, perhaps because it is a complex business. Op-amp ‘accuracy’ is closely related, but the term is often applied only to DC operation. Accuracy here is often specified in terms of bits, so ‘20-bit accuracy’ means errors not exceeding one part in 220, which is −120 dB or 0.0001%. Audio signal distortion is of course a dynamic phenomenon, very sensitive to frequency, and DC specs are of no use at all in estimating it.

Distortion is always expressed as a ratio, and can be quoted as a percentage, as number of decibels, or in parts per million (ppm). With the rise of digital processing, treating distortion as the quantization error arising from the use of a given number of bits has become more popular. Figure 4.2 hopefully provides a way of keeping perspective when dealing with these different metrics.

Figure 4.2: The relation between different ways of quoting THD: decibels, percentages, bit accuracy, and parts per million

There are several different causes of distortion in op-amps. We will now examine them.

Op-Amp Internal Distortion

This is what might be called the basic distortion produced by the op-amp you have selected. Even if you scrupulously avoid clipping, slew-limiting, and common-mode issues, op-amps are not distortion free, though some types such as the 5532 and the LM4562 have very low levels. If distortion appears when the op-amp is run with shunt feedback, to prevent common-mode voltages on the inputs, and with very light output loading, then it is probably wholly internal and there is nothing to be done about it except pick a better op-amp.

If the distortion is higher than expected, the cause may be internal instability provoked by putting a capacitative load directly on the output, or neglecting the supply decoupling. The classic example of the latter effect is the 5532, which shows high distortion if there is not a capacitor across the supply rails close to the package; 100 nF is usually adequate. No actual HF oscillation is visible on the output with a general-purpose oscilloscope, so the problem may be instability in one of the intermediate gain stages.

Slew-Rate-Limiting Distortion

While this is essentially an overload condition, it is wholly the designer’s responsibility. If users whack up the gain until the signal is within a hair of clipping, they should still be able to assume that slew limiting will never occur, even with aggressive material full of high frequencies.

Arranging this is not too much of a problem. If the rails are set at the usual maximum voltage, i.e. ±18 V, then the maximum possible signal amplitude is 12.7 Vrms, ignoring the saturation voltages of the output stage. To reproduce this level cleanly at 20 kHz requires a minimum slew rate of only 2.3 V/μs. Most op-amps can do much better than this, though with the OP27 (2.8 V/μs) you are sailing rather close to the wind. The old LM741 looks as though it would be quite unusable, as its very limited 0.5 V/μs slew rate allows a full output swing only up to 4.4 kHz.

Horrific as it may now appear, audio paths full of LM741s were quite common in the early 1970s. Entire mixers were built with no other active devices, and what complaints there were tended to be about noise rather than distortion. The reason for this is that full-level signals at 20 kHz simply do not occur in reality; the energy at the HF end of the audio spectrum is well known to be much lower than that at the bass end.

This assumes that slew limiting has an abrupt onset as level increases, rather like clipping. This is in general the case. As the input frequency rises and an op-amp gets closer to slew limiting, the input stage is working harder to supply the demands of the compensation capacitance. There is an absolute limit to the amount of current this stage can supply, and when you hit it the distortion shoots up, much as it does when you hit the supply rails and induce voltage clipping. Before you reach this point, the linearity may be degraded, but usually only slightly until you get close to the limit. It is not normally necessary to keep big margins of safety when dealing with slew limiting. If you are employing the usual suspects of the audio op-amp world – the 5532 and TL072, with maximal slew rates of 9 and 13 V/μs respectively – you are most unlikely to suffer any slew-rate non-linearity.

Distortion due to Loading

Output stage distortion is always worse with heavy output loading because the increased currents flowing exacerbate the gain changes in the Class-B output stage. These output stages are not individually trimmed for optimal quiescent conditions (as are audio power amplifiers) and so the crossover distortion produced by op-amps tends to be both higher and more variable between different specimens of the same chip. Distortion increases with loading in different ways for different op-amps. It may rise only at the high-frequency end (e.g. the OP2277) or there may be a general rise at all frequencies. Often both effects occur, as in the TL072.

The lowest load that a given op-amp can be allowed to drive is an important design decision. It will typically be a compromise between the distortion performance required and opposing factors such as number of op-amps in the circuit, cost of load-capable op-amps, and so on. It even affects noise performance, for the lower the load resistance an amplifier can drive, the lower the resistance values in the negative feedback can be, and hence the lower the Johnson noise they generate. There are limits to what can be done in noise reduction by this method, because Johnson noise is proportional to the square root of circuit resistance, and so improves only slowly as op-amp loading is increased.

Thermal Distortion

Thermal distortion is that caused by cyclic variation of the properties of the amplifier components due to the periodic release of heat in the output stage. The result is a rapid rise in distortion at low frequencies, which gets worse as the loading becomes heavier. Those who have read my work on audio power amplifiers will be aware that I am highly sceptical – in fact totally sceptical – about the existence of thermal distortion in amplifiers built from discrete components [3]. The power devices are too massive to experience per-cycle parameter variations, and there is no direct thermal path from the output stage to the input devices. There is no rise, rapid or otherwise, in distortion at low frequencies in a properly designed discrete power amplifier. The situation is quite different in op-amps, where the output transistors have much less thermal inertia and are also on the same substrate as the input devices. Nonetheless, op-amps do not normally suffer from thermal distortion; there is generally no rise in low-frequency distortion, even with heavy output loading. Integrated-circuit power amplifiers are another matter, and the much greater amounts of heat liberated on the substrate do appear to cause serious thermal distortion, rising at 12 dB/octave below 50 Hz. I have never seen anything resembling this in any normal op-amp.

Common-Mode Distortion

This is the general term for extra distortion that appears when there is a large signal voltage on both the op-amp inputs. The voltage difference between these two inputs will be very small, assuming the op-amp is in its linear region, but the common-mode (CM) voltage can be a large proportion of the available swing between the rails.

It appears to be by far the least understood mechanism, and gets little or no attention in op-amp textbooks, but it is actually one of the most important influences on op-amp distortion. It is simple to separate this effect from the basic forward-path distortion by comparing THD performance in series and shunt-feedback modes; this should be done at the same noise gain. The distortion is usually a good deal lower for the shunt-feedback case where there is no common-mode voltage. Bipolar and JFET input op-amps show different behavior and they are treated separately below.

Bipolar Input Op-Amps

Figure 4.4 shows the distortion from a 5532 working in shunt mode with low-value resistors of 1 kΩ and 2k2 setting a gain of 2.2 times, at an output level of 5 Vrms. This is the circuit of Figure 4.3(a) with Rs set to zero; there is no CM voltage. The distortion is well below 0.0005% up to 20 kHz; this underlines what a superlative bargain the 5532 is.

Figure 4.4: 5532 distortion in a shunt-feedback circuit at 5 Vrms out. This shows the AP SYS-2702 output (lower trace) and the op-amp output (upper trace). Supply ±18 V

Figure 4.3: Op-amp test circuits with added source resistance Rs. (a) Shunt. (b) Series. (c) Voltage-follower. (d) Voltage-follower with cancellation resistor in feedback path

Figure 4.5 shows the same situation but with the output increased to 10 Vrms (the clipping level on ±18 V rails is about 12 Vrms) and there is now significant distortion above 10 kHz, though it only exceeds 0.001% at 18 kHz.

Figure 4.5: 5532 distortion in the shunt-feedback circuit of (b). Adding extra resistances of 10 kΩ and 47 kΩ in series with the inverting input does not degrade the distortion at all, but does bring up the noise floor a bit. Test level 10 Vrms out, supply ±18 V

This remains the case when Rs in Figure 4.3(a) is increased to 10 kΩ and 47 kΩ – the noise floor is higher but there is no real change in the audio-band distortion behavior. The significance of this will be seen in a moment.

We will now connect the 5532 in the series-feedback configuration, as in Figure 4.3(b); note that the stage gain is greater at 3.2 times but the op-amp is working at the same noise gain. The CM voltage is 3.1 Vrms. With a 10 Vrms output we can see in Figure 4.6 that even with no added source resistance the distortion starts to rise from 2 kHz, though it does not exceed 0.001% until 12 kHz. But when we add some source resistance Rs, the picture is radically worse, with serious mid-band distortion rising at 6 dB/octave, and roughly proportional to the amount of resistance added. We will note it is 0.0085% at 10 kHz with Rs = 47 kΩ.

Figure 4.6: 5532 distortion in a series-feedback stage with 2k2 and 1k feedback resistors, and varying source resistances. Output 10 Vrms

The worst case for CM distortion is the voltage-follower configuration, as in Figure 4.3(c), where the CM voltage is equal to the output voltage. Figure 4.7 shows that even with a CM voltage of 10 Vrms, the distortion is no greater than for the shunt mode. However, when source resistance is inserted in series with the input, the distortion mixture of second, third, and other low-order harmonics increases markedly. It increases with output level, approximately quadrupling as level doubles. The THD is now 0.018% at 10 kHz with Rs = 47 kΩ, more than twice that of the series-feedback amplifier above, due to the increased CM voltage.

Figure 4.7: 5532 distortion in a voltage-follower circuit with a selection of source resistances. Test level 10 Vrms, supply ±18 V. The lowest trace is the analyzer output measured directly, as a reference

It is would be highly inconvenient to have to stick to the shunt-feedback mode, because of the phase inversion and relatively low input impedance that comes with it, so we need to find out how much source resistance we can live with. Figure 4.8 zooms in on the situation with resistance of 10 kΩ and below; when the source resistance is below 2k2, the distortion is barely distinguishable from the zero source resistance trace. This is why the low-pass Sallen-and-Key filters in Chapter 5 have been given series resistors that do not in total exceed this figure.

Figure 4.8: A closer look at 5532 distortion in a voltage-follower with relatively low source resistances; note that a 1 kΩ source resistance actually gives less distortion than none. Test level 10 Vrms, supply ±18 V

Close examination reveals the intriguing fact that a 1 kΩ source actually gives less distortion than no source resistance at all, reducing THD from 0.00065% to 0.00055% at 10 kHz. Minor resistance variations around 1 kΩ make no difference. This must be due to the cancellation of distortion from two different mechanisms. It is hard to say whether it is repeatable enough to be exploited in practice.

So, what’s going on here? Is it simply due to non-linear currents being drawn by the op-amp inputs? Audio power amplifiers have discrete input stages that are very simple compared with those of most op-amps, and draw relatively large input currents. These currents show appreciable non-linearity even when the output voltage of the amplifier is virtually distortion free, and, if they flow through significant source resistances, will introduce added distortion [4].

If this was the case with the 5532 then the extra distortion would manifest itself whenever the op-amp was fed from a significant source resistance, no matter what the circuit configuration. But we have just seen that it only occurs in series-feedback situations; increasing the source resistance in a shunt-feedback does not perceptibly increase distortion. The effect may be present but if so it is very small, no doubt because op-amp signal input currents are also very small, and it is lost in the noise.

The only difference is that the series circuit has a CM voltage of about 3 Vrms, while the shunt circuit does not, and the conclusion is that with a bipolar input op-amp you must have both a CM voltage and a significant source resistance to see extra distortion. The input stage of a 5532 is a straightforward long-tailed pair (see Figure 4.21 below) with a simple tail-current source, and no fancy cascoding, and I suspect that the Early effect operates on it when there is a large CM voltage, modulating the quite high input bias currents, and this is what causes the distortion. The signal input currents are much smaller, due to the high open-loop gain of the op-amp, and as we have seen appear to have a negligible effect.

Figure 4.21: The internal circuitry of the 5532

FET Op-Amps

FET-input op-amps behave differently from bipolar-input op-amps. Take a look at Figure 4.9, taken from a TL072 working in shunt and in series configuration with a 5 Vrms output. The circuits are as in Figure 4.3(a) and (b), except that the resistor values have to be scaled up to 10 and 22 kΩ because the TL072 is nothing like so good at driving loads as the 5532. This unfortunately means that the inverting input is seeing a source resistance of 10k||22k = 6.9k, which introduces a lot of CM distortion in the series case – five times as much at 20 kHz as for the shunt case. Adding a similar resistance in the input path cancels out this distortion, and the trace then is the same as the ‘Shunt’ trace in Figure 4.9. Disconcertingly, the value that achieved this was not 6.9k, but 9k1. That means adding −113 dBu of Johnson noise, so it’s not always appropriate.

Figure 4.9: A TL072 shunt-feedback stage using 10 and 22 kΩ resistors shows low distortion. The series version is much worse due to the impedance of the NFB network, but it can be made the same as the shunt case by adding cancellation source resistance in the input path. No external loading, test level 5 Vrms, supply ±18 V

It’s worth mentioning that the flat part of the ‘Shunt’ trace below 10 kHz is not noise, as it would be for the 5532; it is distortion.

A voltage-follower has no inconvenient medium-impedance feedback network, but it does have a much larger CM voltage. Figure 4.10 shows a voltage-follower working at 5 Vrms. With no source resistance the distortion is quite low, due to the 100% NFB, but as soon as a 10 kΩ source resistance is added we are looking at 0.015% at 10 kHz.

Figure 4.10: A TL072 voltage-follower working at 5 Vrms with a low source resistance produces little distortion (Rs = 0R), but adding a 10 kΩ source resistance makes things much worse (Rs = 10k). Putting a 10 kΩ resistance in the feedback path as well gives complete cancellation of this extra distortion (Cancel). Supply ±18 V

Once again, this can be cured by inserting an equal resistance in the feedback path of the voltage-follower, as in Figure 4.3(d) above. This gives the ‘Cancel’ trace in Figure 4.10. Adding resistances for distortion cancellation in this way has the obvious disadvantage that they introduce extra Johnson noise into the circuit. Another point is that stages of this kind are often driven from pot wipers, so the source impedance is variable, ranging between zero and one-quarter of the pot track resistance. Setting a balancing impedance in the other op-amp input to a mid-value, i.e. one-eighth of the track resistance, should reduce the average amount of input distortion, but it is inevitably a compromise.

With JFET inputs the problem is not the operating currents of the input devices themselves, which are negligible, but the currents drawn by the non-linear junction capacitances inherent in field-effect devices. These capacitances are effectively connected to one of the supply rails. For P-channel JFETs, as used in the input stages of most JFET op-amps, the important capacitances are between the input JFETs and the substrate, which is normally connected to the V– rail (see Jung [5]).

According to the Burr-Brown data sheet for the OPA2134, ‘The P-channel JFETs in the input stage exhibit a varying input capacitance with applied CM voltage.’ It goes on to recommend that the input impedances should be matched if they are above 2 kΩ.

Common-mode distortion can be minimized by running the op-amp off the highest supply rails permitted, though the differences are not large. In one test on a TL072, going from ±15 to ±18 V rails reduced the distortion from 0.0045% to 0.0035% at 10 kHz.

Rail Bootstrapping to Reduce CM Distortion

So what do you do if you need a really high-impedance low-distortion voltage-follower and you have a significant source resistance, but you don’t want the added noise that would come from adding a cancellation resistor? We noted above that the non-linear input capacitances that cause the trouble with JFET op-amp voltage-followers are effectively connected to the V– supply rail or substrate. This suggests a way to remove the problem: if the supply rails are bootstrapped so they go up and down with the inputs, the signal voltage across the non-linear input capacitances is zero, no current can flow through them, and no extra distortion is generated.

Figure 4.11(a) shows the idea. The resistor–Zener chain R4, D1, D2, R5 creates ±5 V rails that are moved up and down by op-amp A4, and buffered by A2, A3. A1 expects reasonably low supply-rail impedances at HF, and attempting to run it directly from the outputs of A2, A3 does not work – the signal disappears in a fog of HF oscillation. The two resistors R2, R3 prevent this by isolating C1 from A2, A3 outputs, while capacitor C1 across A1 supply pins keeps the HF rail impedance low. Since the ±5 V rails of A1 have to remain inside the fixed ± 15 V supply rails, the possible swing of the supplies is limited and the maximum output is reduced compared with a basic voltage-follower. The circuit of Figure 4.11(a) clips at 6.7 Vrms (1 kHz). This could be increased somewhat by using ±17 or ±18 V fixed rails.

Figure 4.11: Bootstrapping the supply rails of voltage-follower A1 by moving them up and down with the input signal: (a) using op-amps; (b) using transistors

Figure 4.12 shows the result of basic bootstrapping while handling a 5 Vrms signal, which as we saw earlier, is enough to cause serious CM distortion. The increase in linearity is encouraging; the distortion is promptly halved.

Figure 4.12: TL072 voltage-follower distortion with (Y) and without (N) rail bootstrapping. Test level 5 Vrms, supply ±15 V

Figure 4.13, however, shows that we can do better by adding C2, C3. These are in parallel with the effective slope resistance of the Zeners, and improve the accuracy of the rail bootstrapping. The lower trace marked ‘WITH’ is once again indistinguishable from that of the test-gear alone.

Figure 4.13: Rail bootstrapping is much enhanced by adding capacitors C2, C3. TL072, test level 5 Vrms, supply ±15 V. The ‘WITH’ trace is essentially the distortion of the test-gear alone

Figures 4.12 and 4.13 were taken with near-zero source resistance, and show that internal CM distortion has been dealt with. But what happens when a 10 kΩ source resistance is reintroduced?

Figure 4.14 gives the answer: adding a 10 kΩ source resistance now makes almost no difference. Note that no cancellation resistor has been put in the feedback path.

Figure 4.14: TL072 Voltage-follower distortion with 10 kΩ and 50 Ω source resistances, and no cancellation. Test level 3 Vrms, supply ±15 V

Simpler Rail Bootstrapping

On contemplating Figure 4.11(a), it may occur to you that using three op-amps to make a friendly environment for one is a bit over-complex. You are quite right. It is always good to simplicate and add lightness when you can, and A2 and A3 can in fact be replaced by simple emitter-followers with no detectable loss in performance, as in Figure 4.11(b). The two 47 Ω resistors have been removed, but C1 is retained. This seems to be reliably stable. The total supply voltage to A1 has been reduced by two Vbe drops, or 1.2 V; it could be restored by increasing the Zener voltages if required. The simpler version also uses less power as we no longer need to supply the quiescent currents of A2 and A3.

The attentive reader will recall that the troublesome non-linear capacitances are effectively connected to the substrate, which is usually the V– supply rail. Would it not be possible to bootstrap just that rail, and leave V+ connected to a fixed +15 V rail? It would – it works, but the results with an OPA2134, while more linear than with a conventional voltage-follower, are worse than bootstrapping both rails. Before we were keeping the magnitude of the A1 supply voltage substantially constant, although it was sailing up and down. If only one rail is bootstrapped the actual supply voltage is being modulated, so it is hardly surprising that linearity suffers.

The rail bootstrapping concept was also tested with the TL052 and the OPA2134 at 5 Vrms, and similar dramatic reductions in CM distortion were found.

In the previous section we saw that CM distortion is also generated by bipolar input op-amps, though by a different mechanism, so rail bootstrapping ought to work for these types of op-amp as well. Figure 4.15 shows that it does. Adding a 10 kΩ source resistance now causes virtually no extra distortion.

Figure 4.15: Rail bootstrapping works for 5532 voltage-followers as well; 10 kΩ and 50 Ω source resistances, and no cancellation. Test level 5 Vrms, supply ±15 V

Bootstrapping Series-Feedback JFET Op-Amp Stages

The voltage-follower is the worst case for CM distortion, as the full output voltage exists on both inputs. In contrast, it is the best case for output loading, as there is no resistive feedback network at all to drive – just a high-impedance input pin. Similarly, the shunt-feedback amplifier is the best case for CM distortion as there is no significant signal on the inputs.

Series-feedback amplifier stages fall between these two cases. For a +10 dB amplifier stage, the signal on the inputs is one-third that of the output, and so the input distortion is less, but still very definitely present, as we saw in Figures 4.6 and 4.9. Amplifier stages like this can have a mixture of distortion mechanisms. The impedance of the NFB network, as seen from the inverting input of the amplifiers, is 22 kΩ in parallel with 10 kΩ, i.e. 6.87 kΩ. We have seen above that this is enough to cause serious non-linearity unless the other input sees the same impedance, and it might be thought that reducing the impedance level of the NFB network would be a good way to deal with this, not least because it would minimize the Johnson noise produced by the network. Figure 4.16 shows that this does not work for the TL072; if the feedback network impedance is reduced by a factor of 10 the distortion gets worse rather than better, due to the heavier loading on the output. Input distortion has been replaced by a larger amount of output distortion; this is not a good exchange. Lowering the NFB network impedance is, however, likely to be successful with JFET op-amps having better load-driving capability than the TL072.

Figure 4.16: With the TL072, reducing the impedance of the negative-feedback network may reduce input distortion, but output distortion more than makes up for it because of the extra loading. Upper trace 2k2 – 1 kΩ, lower trace 22 kΩ – 10 kΩ in feedback network

Rail bootstrapping is once more a possible answer. We drive the op-amp supply rails up and down with the same signal as the input – not the output. The only modification required is to take the increased output swing into account by increasing the A1 supply voltage to ±10 V (see Figure 4.17). The Zeners have been replaced with simple resistive dividers. This works just as well, and is a good thing as Zeners are more expensive than resistors. Figure 4.18 shows the excellent results.

Figure 4.17: Bootstrapping the rails of a series-feedback amplifier from the input of A1. The Zeners have been replaced by resistors

Figure 4.18: The benefit of bootstrapping the rails of a series-feedback amplifier with a gain of 3.2×. The lower trace is essentially that of the THD from the test equipment

Selecting the Right Op-Amp

In audio work, the 5532 is pre-eminent. It is found in almost every mixing console, and in a large number of preamplifiers. Distortion is almost unmeasurably low, even when driving 600 Ω loads. Noise is very low, and the balance of voltage and current noise in the input stage is well matched to moving-magnet phono cartridges; in many applications discrete devices give no significant advantage. Large-quantity production has brought the price down to a point where a powerful reason is required to pick any other device.

The 5532 is not, however, perfect. It suffers common-mode distortion. It has high bias and offset currents at the inputs, as an inevitable result of using a bipolar input stage (for low noise) without any sort of bias-cancellation circuitry. The 5532 is not in the forefront for DC accuracy, though it’s not actually that bad. The offset voltage spec is 0.5 mV typical, 4 mV max, compared with 3 mV typical, 6 mV max for the popular TL072. I have actually used 5532s to replace TL072s when offset voltage was a problem, but the increased bias current was acceptable.

With horrible inevitability, the very popularity and excellent technical performance of the 5532 has led to it being criticized by subjectivists who have contrived to convince themselves that they can tell op-amps apart by listening to music played through them. This always makes me laugh, because there is probably no music on the planet that has not passed through a hundred or more 5532s on its way to the consumer.

In some applications, such as low-cost mixing consoles, bipolar-style bias currents are a real nuisance because keeping them out of EQ pots to prevent scratching noises requires an inappropriate number of blocking capacitors. There are plenty of JFET-input op-amps around with negligible bias currents, but there is no obviously superior device that is the equivalent of the 5532. The TL072 has been used in this application for many years but its HF linearity is not first-class and distortion across the band deteriorates badly as output loading increases. However, the op-amps in many EQ sections work in the shunt-feedback configuration with no CM voltage on the inputs, and this reduces the distortion considerably. When low bias currents are needed with superior performance then the OPA2134 is often a good choice, though it is at least four times as expensive as the TL072.

The question of op-amp selection is examined in much more detail in the rest of this chapter, where the most popular types are surveyed.

Op-Amps Surveyed: BJT Input Types

The rest of this chapter looks at some op-amp types and examines their performance, with the 5532 the usual basis for comparison. The parts shown here are not necessarily intended as audio op-amps, though some, such as the OP275 and the OPA2134, were specifically designed as such. They have, however, all seen use, in varying numbers, in audio applications. Bipolar input op-amps are dealt with first.

The LM741 Op-Amp

The LM741 is only included here for its historical interest; in its day it was a most significant development and, to my mind, the first really practical op-amp. It was introduced by Fairchild in 1968 and is considered a second-generation op-amp, the 709 being first generation.

The LM741 had (and indeed has) effective short-circuit protection and internal compensation for stability at unity gain, and was much easier to make work in a real circuit than its predecessors. It was clear that it was noisy compared with discrete circuitry, and you sometimes had to keep the output level down if slew limiting was to be avoided, but with care it was usable in audio. Probably the last place the LM741 lingered was in the integrators of state-variable EQ filters, where neither indifferent noise performance nor poor slewing capability is a serious problem (see Chapter 10 for more details on this application). The LM741 is a single op-amp. The dual version is the LM747.

Figure 4.19 shows a region between 100 Hz and 4 kHz where distortion rises at 6dB/octave. This is the result of the usual dominant-pole Miller compensation scheme. When slew limiting begins, the slope increases and THD rises rapidly with frequency.

Figure 4.19: The THD performance of an LM741 working at a gain of 3×, on ±15 V rails, giving 3 and 6 Vrms outputs, with no load. At 6 Vrms, slew distortion exceeds 1% before 20 kHz is reached; there is visible slew limiting in the waveform. THD is, however, very low at 100 Hz, due to the high NFB factor at low frequencies

The NE5532/5534 Op-Amp

The 5532 is a low-noise, low-distortion bipolar dual op-amp, with internal compensation for unity-gain stability. The 5534 is a single version internally compensated for gains down to 3, and an external compensation capacitor can be added for unity-gain stability; 22 pF is the usual value. The common-mode range of the inputs is a healthy ±13 V, with no phase inversion problems if this is exceeded. The 5532 has a distinctly higher power consumption than the TL072, drawing approximately 4 mA per op-amp section when quiescent. The DIL version runs perceptibly warm when quiescent on ±17 V rails.

Figure 4.20 shows that the 5532 deals well with loads up to its maximum 500 Ω. Its distortion performance is studied in detail in the section above on common-mode distortion.

Figure 4.20: Distortion is very low from the 5532, though loading makes a detectable difference. Here it is working in series feedback mode at the high level of 10 Vrms with 500 Ω, 1 kΩ loads and no load. The ‘Gen-mon’ trace is the output of the distortion analyzer measured directly. Gain of 3.2×, supply ±18 V

The 5534/5532 has bipolar transistor input devices. This means it gives low noise with low source resistances, but draws a relatively high bias current through the input pins. The input devices are NPN, so the bias currents flow into the chip from the positive rail. If an input is fed through a significant resistance then the input pin will be more negative than ground due to the voltage drop caused by the bias current. The inputs are connected together with back-to-back diodes for reverse-voltage protection, and should not be forcibly pulled to different voltages. The 5532 is intended for linear operation, and using it as a comparator is not recommended.

As can be seen from Figure 4.20, the 5532 is almost distortion free, even when driving the maximum 500 Ω load. The internal circuitry of the 5532 has never been publicly explained, but appears to consist of nested Miller loops that permit high levels of internal negative feedback. The 5532 is the dual of the 5534, and is more commonly used than the single as it is cheaper per op-amp and does not require an external compensation capacitor when used at unity gain.

The 5532 and 5534 type op-amps require adequate supply decoupling if they are to remain stable, otherwise they appear to be subject to some sort of internal oscillation that degrades linearity without being visible on a normal oscilloscope. The essential requirement is that the positive and negative rails should be decoupled with a 100 nF capacitor between them, at a distance of not more than a few millimeters from the op-amp; normally one such capacitor is fitted per package as close to it as possible. It is not necessary, and often not desirable, to have two capacitors going to ground; every capacitor between a supply rail and ground carries the risk of injecting rail noise into the ground.

Deconstructing the 5532

To the best of my knowledge, virtually nothing has been published about the internal operation of the 5532. This is surprising given its unique usefulness as a high-quality audio op-amp. I believe the secret of the 5532’s superb linearity is the use of nested negative feedback inside the circuit, in the form of traditional Miller compensation.

Figure 4.21 shows the only diagram of the internal circuitry that has been released; the component and node numbers are mine. This has been in the public domain for at least 20 years, so I hope no one is going to object to my impertinent comments on it. The circuit initially looks like a confusing sea of transistors, and there is even a solitary JFET lurking in there, but it breaks down fairly easily. There are three voltage-gain stages, plus a unity-gain output stage to increase drive capability. This has current-sensing overload protection. There is also a fairly complex bias generator that establishes the operating currents in the various stages.

In all conventional op-amps there are two differential input signals that have to be subtracted to create a single output signal, and the node at which this occurs is called the ‘phase-summing point’.

Q1, Q2 make up the input differential amplifier. They are protected against reverse biasing by the diode-connected transistors across the input pins. Note there are no emitter degeneration resistors, which would linearize the input pair at the expense of degrading noise. Presumably high open-loop gain (note there are three gain stages, whereas a power amplifier normally only has two) means that the input pair is handling very small signal levels, so its distortion is not a problem.

Q3, Q4 make up the second differential amplifier; emitter degeneration is now present. Phase summing occurs at the output of this stage at node 2. C1 is the Miller capacitor around this stage, from node 2 to node 1. Q5, Q6, Q7 are a Wilson current mirror, which provides a driven current source as the collector load of Q4. The function of C4 is obscure but it appears to balance C1 in some way.

The third voltage-amplifier stage is basically Q9 with split-collector transistor Q15 as its current-source load. Q8 increases the basic transconductance of the stage, and C3 is the Miller capacitor around it, feeding from node 3 to node 2 – note that this Miller loop does not include the output stage. Things are a bit more complicated here as it appears that Q9 is also the sink half of the Class-B output stage. Q14 looks very mysterious as it seems to be sending the output of the third stage back to the input; possibly it’s some sort of clamp to ensure clean clipping, but to be honest I haven’t a clue. Q10 plus associated diode generates the bias for the Class-B output stage, just as in a power amplifier.

The most interesting signal path is the semi-local Miller loop through C2, from node 3 to node 1, which encloses both the second and third voltage amplifiers; each of these has its own local Miller feedback, so there are two nested layers of internal feedback. This is probably the secret of the 5532’s low distortion.

Q11 is the source side of the output stage and, as mentioned above, Q9 appears to be the sink. Q12, Q13 implement overcurrent protection. When the voltage drop across the 15 Ω resistor becomes too great, Q12 turns on and shunts base drive away from Q11. In the negative half-cycle, Q13 is turned on, which in turn activates Q17 to shunt drive away from Q8.

The biasing circuit shows an interesting point. Bipolar bias circuits tend not to be self-starting; no current flows anywhere until some flows somewhere, so to speak. Relying on leakage currents for starting is unwise, so here the depletion-mode JFET provides a circuit element that is fully on until you bias it off, and can be relied upon to conduct as the power rails come up from zero.

The LM4562 Op-Amp

The LM4562 is a new op-amp, which first became freely available at the beginning of 2007. It is a National Semiconductor product. It is a dual op-amp – there is no single or quad version. It costs about 10 times as much as a 5532.

The input noise voltage is typically image, which is substantially lower than the image image of the 5532. For suitable applications with low source impedances this translates into a useful noise advantage of 3.4 dB. The bias current is 10 nA typical, which is very low and would normally imply that bias cancellation, with its attendant noise problems, was being used. However, in my testing I have seen no sign of excess noise, and the data sheet is silent on the subject. No details of the internal circuitry have been released so far, and quite probably never will be.

It is not fussy about decoupling and, as with the 5532, 100 nF across the supply rails close to the package should ensure HF stability. The slew rate is typically ±20 V/μs, more than twice as quick as the 5532.

The first THD plot in Figure 4.22 shows the LM4562 working at a closed-loop gain of 2.2× in shunt-feedback mode, at a high level of 10 Vrms. The top of the THD scale is 0.001%, something you will see with no other op-amp in this survey. The no-load trace is barely distinguishable from the AP SYS-2702 output, and even with a heavy 500 Ω load driven at 10 Vrms there is only a very small amount of extra THD, reaching 0.0007% at 20 kHz.

Figure 4.22: The LM4562 in shunt-feedback mode, with 1 kΩ, 2k2 feedback resistors giving a gain of 2.2×. Shown for no load (NL) and 1 kΩ, 500 Ω loads. Note the vertical scale ends at 0.001% this time. Output level is 10 Vrms, ±18 V supply rails

Figure 4.23 shows the LM4562 working at a gain of 3.2× in series-feedback mode, both modes having a noise gain of 3.2×. The extra distortion from 500 Ω loading is barely detectable.

Figure 4.23: The LM4562 in series-feedback mode, with 1 kΩ, 2k2 feedback resistors giving a gain of 3.2×. No load (NL) and 500 Ω load. Output 10 Vrms, ±18 V supply rails

For Figures 4.23 and 4.24 the feedback resistances were 2k2 and 1 kΩ, so the minimum source resistance presented to the inverting input is 687 Ω. In Figure 4.25 extra source resistances were then put in series with the input path (as was done with the 5532 in the section above on common-mode distortion) and this revealed a remarkable property of the LM4562 – it is much more resistant to common-mode distortion than the 5532. At 10 Vrms and 10 kHz, with a 10 kΩ source resistance the 5532 generates 0.0014% THD (see Figure 4.6) but the LM4562 gives only 0.00046% under the same conditions. I strongly suspect that the LM4562 has a more sophisticated input stage than the 5532, probably incorporating cascoding to minimize the effects of common-mode voltages.

Figure 4.24: The LM4562 in series-feedback mode, gain 3.2×, with varying extra source resistance in the input path. The extra distortion is much lower than for the 5532. Output 10 Vrms, ±18 V supply rails

Figure 4.25: AD797 THD into loads down to 500 Ω, at 7.75 Vrms. Output is virtually indistinguishable from input. Series feedback, but no CM problems. Gain = 3.2×

Note that only the rising curves to the right represent actual distortion. The raised levels of the horizontal traces at the LF end are due to Johnson noise from the extra series resistance.

It has taken an unbelievably long time – nearly 30 years – for a better audio op-amp than the 5532 to come along, but at last it has happened. The LM4562 is superior in just about every parameter, but it has much higher current noise. At present it also has a much higher price, but hopefully that will change.

The AD797 Op-Amp

The AD797 (Analog Devices) is a single op-amp with very low voltage noise and distortion. It appears to have been developed primarily for the cost-no-object application of submarine sonar, but it works very effectively with normal audio – if you can afford to use it. The cost is something like 20 times that of a 5532. No dual version is available, so the cost ratio per op-amp section is 40 times.

This is a remarkably quiet device in terms of voltage noise, but current noise is correspondingly high due to the high currents in the input devices. Early versions appeared to be rather difficult to stabilize at HF, but the current product is no harder to apply than the 5532. Possibly there has been a design tweak, or on the other hand my impression may be wholly mistaken.

The AD797 incorporates an ingenious feature for internal distortion cancellation. This is described on the manufacturer’s data sheet. Figure 4.25 shows that it works effectively.

The OP27 Op-Amp

The OP27 from Analog Devices is a bipolar-input single op-amp primarily designed for low noise and DC precision. It was not intended for audio use, but in spite of this it is frequently recommended for such applications as RIAA and tape head preamps. This is unfortunate, because while at first sight it appears that the OP27 is quieter than the 5534/5532, as the en is image compared with image for the 5534, in practice it is usually slightly noisier.

This is because the OP27 is in fact optimized for DC characteristics, and so has input bias-current cancellation circuitry that generates common-mode noise. When the impedances on the two inputs are very different – which is the case in RIAA preamps – the CM noise does not cancel, and this appears to degrade the overall noise performance significantly.

For a bipolar input op-amp, there appears to be a high-level common-mode input distortion, enough to bury the output distortion caused by loading (see Figures 4.26 and 4.27). It is likely that this too is related to the bias-cancellation circuitry, as it does not occur in the 5532.

Figure 4.26: OP27 THD in shunt-feedback mode with varying loads. This op-amp accepts even heavy (1 kΩ) loading gracefully

Figure 4.27: OP27 THD in series-feedback mode. The common-mode input distortion completely obscures the output distortion

The maximum slew rate is low compared with other op-amps, being typically 2.8 V/μs. However, this is not the problem it may appear. This slew rate would allow a maximum amplitude at 20 kHz of 16 Vrms, if the supply rails permitted it. I have never encountered any particular difficulties with decoupling or stability of the OP27.

The OP270 Op-Amp

The OP270 from Analog Devices is a dual op-amp, intended as a ‘very-low-noise precision operational amplifier’, in other words combining low noise with great DC accuracy. The input offset voltage is an impressive 75 μV maximum. It has bipolar inputs with a bias-current cancellation system; the presence of this is shown by the 15 nA bias current spec, which is 30 times less than the 500 nA taken by the 5534, which lacks this feature. It will degrade the noise performance with unequal source resistances, as it does in the OP27. The input transistors are protected by back-to-back diodes.

The OP270 distortion performance suffers badly when driving even modest loads. See Figures 4.28 and 4.29. The slew rate is a rather limited 2.4 V/μs, which is only just enough for a full output swing at 20 kHz. Note also that this is an expensive op-amp, costing something like 25 times as much as a 5532; precision costs money. Unless you have a real need for DC accuracy, this part is not recommended.

Figure 4.28: OP270 THD in shunt-feedback mode. Linearity is severely degraded even with a 2k2 load

Figure 4.29: OP270 THD in series-feedback mode. This looks the same as in so CM input distortion appears to be absent

The OP275 Op-Amp

The Analog Devices OP275 is one of the few op-amps specifically marketed as an audio device. Its most interesting characteristic is the Butler input stage, which combines bipolar and JFET devices. The idea is that the bipolars give accuracy and low noise, while the JFETs give speed and ‘the sound quality of JFETs’. That final phrase is not a happy thing to see on a data sheet from a major manufacturer; the sound of JFETs (if any) would be the sound of high distortion. Just give us the facts, please.

The OP275 is a dual op-amp; no single version is available. It is quite expensive, about six times the price of a 5532, and its performance in most respects is inferior. It is noisier, has higher distortion, and does not like heavy loads. See Figures 4.30 and 4.31. The CM range is only about two-thirds of the voltage between the supply rails, and Ibias is high due to the BJT part of the input stage. Unless you think there is something magical about the BJT/JFET input stage – and I am quite sure there is not – it is probably best avoided.

Figure 4.30: An OP275 driving 7.75 Vrms into no load and 600 Ω. THD below 1 kHz is definitely non-zero with the 600 Ω load. Series feedback, gain 3.2×

Figure 4.31: OP275 driving 5 Vrms into 1 k and 600 Hz Ω. Shunt feedback, gain 2.2×, but note noise gain was set to 3.2× as for the series case. The ‘Gen-Mon’ trace shows the distortion of the AP System 2 generator; the steps at 200 Hz and 20 kHz are artefacts generated by internal range switching

The THD at 10 kHz with a 600 Ω load is 0.0025% for shunt and 0.009% for series feedback; there is significant CM distortion in the input stage, which is almost certainly coming from the JFETs. (I appreciate the output levels are not the same but I think this only accounts for a small part of the THD difference.) Far from adding magical properties to the input stage, the JFETs seem to be just making it worse.

Op-Amps Surveyed: JFET Input Types

Op-amps with JFET inputs tend to have higher voltage noise and lower current noise than BJT-input types, and therefore give a better noise performance with high source resistances. Their very low bias currents often allow circuitry to be simplified.

The TL072 Op-Amp

The TL072 is one of the most popular op-amps, having very-high-impedance inputs, with effectively zero bias and offset currents. The JFET input devices give their best noise performance at medium impedances, in the range 1–10 kΩ. The TL072 has a modest power consumption at typically 1.4 mA per op-amp section, which is significantly less than with the 5532. The slew rate is higher than for the 5532, at 13 V/μs against 9 V/μs. The TL072 is a dual op-amp. There is a single version called the TL071, which has offset null pins.

However, the TL072 is not THD free in the way the 5532 is. In audio usage, distortion depends primarily upon how heavily the output is loaded. The maximum loading is a trade-off between quality and circuit economy, and I would put 2 kΩ as the lower limit. This op-amp is not the first choice for audio use unless the near-zero bias currents (which allow circuit economies by making blocking capacitors unnecessary), the low price, or the modest power consumption are dominant factors.

It is a quirk of this device that the input common-mode range does not extend all the way between the rails. If the common-mode voltage gets to within a couple of volts of the V – rail, the op-amp suffers phase reversal and the inputs swap their polarities. There may be really horrible clipping, where the output hits the bottom rail and then shoots up to hit the top one, or the stage may simply latch up until the power is turned off.

TL072s are relatively relaxed about supply-rail decoupling, though they will sometimes show very visible oscillation if they are at the end of long thin supply tracks. One or two rail-to-rail decoupling capacitors (e.g. 100 nF) per few centimeters is usually sufficient to deal with this, but normal practice is to not take chances, and allow one capacitor per package as with other op-amps.

Because of common-mode distortion, a TL072 in shunt configuration is always more linear. In particular compare the results for 3k3 load in Figures 4.32 and 4.33. At heavier loadings the difference is barely visible because most of the distortion is coming from the output stage.

Figure 4.32: Distortion versus loading for the TL072, with various loads. Shunt-feedback configuration eliminates CM input distortion. Output level 3 Vrms, gain 3.2×, rails ±15 V. No output load except for the feedback resistor. The no-load plot is indistinguishable from that of the test-gear alone. Distortion always gets worse as the loading increases. This factor, together with the closed-loop NFB factor, determines the THD

Figure 4.33: Distortion versus loading for the TL072, with various loads. Series-feedback configuration, output level 3 Vrms, gain 3.2×, rails ±15 V. Distortion at 10 kHz with no load is 0.0015% compared with 0.0010% for the shunt configuration. This is due to the 1 Vrms CM signal on the inputs

TL072/71 op-amps are prone to HF oscillation if faced with significant capacitance to ground on the output pin; this is particularly likely when they are used as unity-gain buffers with 100% feedback. A few inches of track can sometimes be enough. This can be cured by an isolating resistor, in the 47–75 Ω range, in series with the output, placed at the op-amp end of the track.

The TL052 Op-Amp

The TL052 from Texas Instruments was designed to be an enhancement of the TL072, and so is naturally compared with it. Most of the improvements are in the DC specifications. The offset voltage is 0.65 mV typical, 1.5 mV max, compared with the TL072’s 3 mV typical, 10 mV max. It has half the bias current of the TL072. This is very praiseworthy, but rarely of much relevance to audio.

The distortion, however, is important, and this is worse rather than better. THD performance is rather disappointing. The unloaded THD is low, as shown in Figure 4.34, in series-feedback mode. As usual, practical distortion depends very much on how heavily the output is loaded. Figure 4.35 shows that it deteriorates badly for loads of less than 4k7.

Figure 4.34: Distortion versus frequency at two output levels for the TL052CP, with no load. Series feedback

Figure 4.35: Distortion of the TL052 at 5 Vrms output with various loads. At 1 kΩ and 2k2 loading the residual is all crossover distortion at 1 kHz. Gain 3.2×, non-inverting (series feedback)

The slew rate is higher than for the TL072 (18 against 13 V/μs) but the lower figure is more than adequate for a full-range output at 20 kHz, so this enhancement is of limited interest. The power consumption is higher, typically 2.3 mA per op-amp section, which is almost twice that of the TL072. Like the TL072, the TL052 is relatively relaxed about supply-rail decoupling. At the time of writing (2009) the TL052 costs at least twice as much as the TL072.

The OPA2134 Op-Amp

The OPA2134 is a Burr-Brown product, the dual version of the OPA134. The manufacturer claims it has superior sound quality, due to its JFET input stage. Regrettably, but not surprisingly, no evidence is given to back up this assertion. The input noise voltage is image image, almost twice that of the 5532. The slew rate is typically ±20 V/μs, which is ample. The OPA2134 does not appear to be optimized for DC precision, the typical offset voltage being ±1 mV, but this is usually good enough for audio work. I have used it many times as a DC servo in power amplifiers, the low bias currents allowing high resistor values and correspondingly small capacitors.

The OPA2134 does not show phase reversal anywhere in the common-mode range, which immediately marks it as superior to the TL072.

The two THD plots in Figures 4.36 and 4.37 show the device working at a gain of 3× in both shunt and series-feedback modes. It is obvious that a problem emerges in the series plot, where the THD is higher by about three times at 5 Vrms and 10 kHz. This distortion increases with level, which immediately suggests common-mode distortion in the input stage. Distortion increases with even moderate loading, see Figure 4.38.

Figure 4.36: The OPA2134 working in shunt-feedback mode. The THD is below the noise until frequency reaches 10 kHz; it appears to be lower at 5 Vrms simply because the noise floor is relatively lower.

Figure 4.37: The OPA2134 in series-feedback mode. Note much higher distortion at HF

Figure 4.38: The OPA2134 in shunt-feedback mode (to remove input CM distortion) and with varying loads on the output. As usual, more loading makes linearity worse. Output 5 Vrms, gain = 3.3×

This is a relatively modern and sophisticated op-amp. When you need JFET inputs (usually because significant input bias currents would be a problem) this definitely beats the TL072; it is, however, four to five times more expensive.

The OPA604 Op-Amp

The OPA604 from Burr-Brown is a single JFET-input op-amp, which claims to be specially designed to give low distortion. The simplified internal circuit diagram in the data sheet includes an enigmatic box intriguingly labeled ‘Distortion Rejection Circuitry’. This apparently ‘linearizes the open-loop response and increases voltage gain’, but no details as to how are given; whatever is in there appears to have been patented so it ought to be possible to track it down. However, despite this, the distortion is not very low even with no load (see Figure 4.39), and is markedly inferior to the 5532’s. The OPA604 is not optimized for DC precision, the typical offset voltage being ±1 mV. The OPA2604 is the dual version, which omits the offset null pins.

Figure 4.39: An OPA2604 driving various loads at 7.75 Vrms. Series feedback, gain = 3.2×

The data sheet includes a discussion that attempts to show that JFET inputs produce a more pleasant type of distortion than BJT inputs. This unaccountably omits the fact that the much higher transconductance of BJTs means that they can be linearized by emitter degeneration so that they produce far less distortion of whatever type than a JFET input [6]. Given that the OPA604 costs five times as much as a 5532, it is not very clear under what circumstances this op-amp would be a good choice.

The OPA627 Op-Amp

The OPA627 from Burr-Brown is a laser-trimmed JFET-input op-amp with excellent DC precision, the input offset voltage being typically ±100 μV. The distortion is very low, even into a 600 Ω load, though it is increased by the usual common-mode distortion when series feedback is used.

The OPA627 is a single op-amp and no dual version is available. The OPA637 is a decompensated version only stable for closed-loop gains of 5 or more. This op-amp makes a brilliant DC servo for power amplifiers, if you can afford it; it costs about 50 times as much as a 5532, which is 100 times more per op-amp section, and about 20 times more per op-amp than the OPA2134, which is my usual choice for DC servo work.

The current noise in is very low, the lowest of any op-amp examined in this book, apparently due to the use of Difet (dielectrically isolated JFET) input devices, and so it will give a good noise performance with high source resistances. Voltage noise is also very respectable at image, only fractionally more than that of the 5532.

The series-feedback case barely has more distortion than the shunt one, and only at the extreme HF end. It appears that the Difet input technology also works well to prevent input non-linearity and CM distortion. See Figures 4.40 and 4.41.

Figure 4.40: OPA627 driving the usual loads at 5 Vrms. Series feedback, gain = 3.2×. ‘Gen-Mon’ is the test-gear output

Figure 4.41: OPA627 driving the usual loads at 5 Vrms. Shunt feedback, gain = 2.2× but noise gain = 3.2×. ‘Gen-Mon’ trace shows the distortion produced by the AP System 2 generator alone

References

[1]  A. Blumlein, UK patent 482,470, 1936.

[2]  W. Jung (Ed.), Op-Amp Applications Handbook, Newnes, 2006 (Chapter 8).

[3]  D. Self, Audio Power Amplifier Design Handbook, fifth ed, Focal Press, 2009, pp. 186–189.

[4]  D. Self, Audio Power Amplifier Design Handbook, fifth ed, Focal Press, 2009, p. 96.

[5]  W. Jung (Ed.), Op-Amp Applications Handbook, Newnes, 2006, p. 399 (Chapter 5).

[6]  D. Self, Audio Power Amplifier Design Handbook, fifth ed, Focal Press, 2009, p. 380.

Small Signal Audio Design; ISBN: 9780240521770

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