Chapter 4

Receiving Amplifying Antennas

4.1 Introduction

As early as the 1960s, dipoles were integrated with low-noise amplifiers (LNAs) to enhance the signal-to-noise ratio (SNR) at receivers (1–3). The new integrated antenna was called an antennafier because it combined the functions of radiation and amplification. Such integration was used to reduce the number of matching and tuning elements for a lower RF loss. When integrated with a transmitter (3), the power amplifying antenna was also called an antennamitter. In 1997, Radisic et al. (4) used a patch antenna as the harmonic tuning load of a power amplifier. By terminating the second harmonic of the power amplifier, the power-added efficiency (PAE) was found to be better. This method was later extended for suppressing the higher harmonics of a push–pull power amplifier (5). Again, significant enhancement (with PAE > 55%) in the output power was observed. In the past two decades, antennas have been combined with both transmitters and receivers for designs of multifunctional transceivers (6, 7). Also, a lot of research has been reported on the integration of antenna and transceiver for quasi-optical spatial power combining during the millimeter-wave technology boom (8–11).

Obviously, the main advantage of integrating an antenna with an amplifier is the significant reduction in the length of signal transmission path that leads to improvement of signal quality and reduction of circuit size. Although fewer circuit elements are used, the impedance matching for the amplifying antennas remains very important. Since power amplifying antennas have been covered extensively by many publications and books (e.g., (1–11)), in this chapter we focus on discussing the matching techniques and new applications of low-noise amplifying antennas. Co-design and co-optimization processes of such active antennas are also explored.

4.2 Design Criteria and Considerations

Figure 4.1 shows the functional blocks of an LNA, including its input- and output-matching circuits, at a typical receiver. Good stability, high gain, low noise, low power consumption, and low cost are among the important criteria of an LNA. To have unconditional stability, in amplifier design, it is crucial to bias a transistor so that the conditions stated in (4.1) and (4.2) are met. Condition (4.2) implies (4.3) and (4.4).

4.1

4.2

4.3

4.4

For an LNA, it is always very desirable to have a low noise factor F, which is defined as the ratio of the SNR at the input to the SNR at the output. The noise factor in decibels (10 log₁₀F) is also called the noise figure (NF). Obviously, it is directly proportional to the noise that a transistor adds to the output signal. For a typical receiving active amplifying antenna, as shown in Fig. 4.2, the input-matching circuit is usually not needed. This can significantly reduce the footprint and circuit size.

Figure 4.1 Block diagram of a low-noise amplifier at a typical receiver.

Figure 4.2 Block diagram of a receiving active amplifying antenna.

4.3 Wearable Low-noise Amplifying Antenna

Configuration

In recent years, many passive wearable antennas have been reported (12–16). Research was also conducted to study the effects of the human body on such antennas (17, 18). In 2010, Declercq and Rogier (19) first proposed an active wearable antenna, made on garments. To demonstrate the design idea, an LNA is connected directly to a textile antenna without using an input-matching network. The impedance of the antenna is designed to match the input impedance of the LNA for optimal noise performance. In this case, the antenna impedance is designed to be coincident with the optimal impedance (of the transistor) for achieving a minimum noise figure. The schematic of the amplifier is shown in Fig. 4.3. It is designed using the biasing scheme recommended by Avago Technologies (20). With the use of the power dividing resistors R₁ = 300 Ω, R₂ = 1.2 kΩ, and R₃ = 10 Ω, an Avago ATF-54143 + e-PHEMT transistor is biased in the common-source configuration with V_ds = 3 V, I_d = 60 mA, and V_gs = 0.56 V. A high-value resistor R₄ = 10 kΩ has been used as the current limiter of the gate. Resistor R₅ = 50 Ω functions as a resistive termination so that the low-frequency stability can be enhanced. The inductor L₁ = 2.7 nH and the λ/4 high-impedance (100-Ω) microstrip are both RF chokes, while C₃ is a dc block. The components C₁ = C₄ = 8.2 pF and C₂ = 10, 000 pF are bypass capacitors, where C₂ can further improve the low-frequency stability. An inductor L₂ is for output matching, while R₆ is for resistive loading. A flexible substrate made of a woven textile layer and a very thin flexible polyimid layer is used as a circuit board for accommodating the active devices and interconnects. The input port is connected directly to the receiving antenna through a via (diameter of 1 mm) to achieve low loss, low noise, and compact size.

Figure 4.3 Schematic of an LNA (19).

Figure 4.4 shows the configuration of the ring antenna being used. It is built on a double-layered substrate and is designed to cover the entire 2.4 to 2.484 GHz ISM band. The top layer is made of a flexible polyurethane foam with a thickness of h₁ = 3.56 mm. As can be seen from the figure, the bottom layer consists of an aramid fabric (400 μm) and a polyimid textile (25 μm), together totaling to a substrate thickness of h₂ = 425 μm. The antenna and ground are made from a type of copper-plated nylon fabric called Flectron. All the materials are glued together by adhesive. The properties of the materials are given in Table 4.1. An active wearable antenna that has a total area of 11 cm² is shown in Fig. 4.5. As can be seen from the figure, the polyimid layer is kept as small as possible. Good bonding is formed between the aramid and polyimid layers.

Figure 4.4 Configuration of an active wearable antenna: (a) top view; (b) front view.

Figure 4.5 Active wearable antenna: (a) bottom view; (b) top view. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

Table 4.1 Properties of the Materials for an Active Wearable Antenna

The co-design and co-optimization processes of the antenna and amplifier require both field- and SPICE-based CAD softwares. In this case, the commercial software Agilent ADS-Momentum and CST Microwave Studio are used jointly in designing the LNA and antenna, respectively. Material properties given in Table 4.1 have been used for all the simulations. To begin with, the LNA design is discussed. The transistor and all the lumped elements are simulated using Agilent's ADS. Momentum, a full-wave software affiliated with the ADS, can be employed for simulating the passive interconnects placed on the polyimid so that the parasitic effect can be accounted for. Then the antenna can be simulated using the field-based simulation software CST Microwave Studio, and the components can be combined using the following three-step co-design and co-optimization strategy (19):

As many solutions may meet the requirements, the simulated annealing was used as an optimization algorithm to find the global maximum. The co-optimized antenna and LNA parameters are shown in Tables 4.2 and 4.3. Also given in the tables are the design parameters for the isolated antenna and LNA. In general, the two groups of parameters are quite close to each other.

Source: (19).

Table 4.2 Design Parameters of an Isolated and Co-optimized Antenna

Source: (19).

Table 4.3 Design Parameters (Millimeters) of an Isolated and Co-optimized LNA

4.3.1.1 Results and Discussion

LNA Performance

The amplifier is first designed on the fabric materials without including the ring antenna. To measure the LNA output, the antenna is removed and the amplifier output is soldered to the center conductor of an SMA connector punching through the fabric. An Agilent N5242A PNA-X vector network analyzer (VNA) is used to measure the S parameters. The source-corrected noise measurement technique, which is a function affiliated to the PNA-X VNA, is adopted for measuring the noise figures (21). To account for the effects of the human body, the LNA is placed on a person's back for measurements. With reference to Fig. 4.6, the simulated and measured amplifier gains (|S₂₁|) and noise figures are compared, with good agreement observed. The measured amplifier gain is 11.18 dB at 2.45 GHz, slightly lower than that for simulation (11.79 dB). As can be seen from the figure, the measured noise figure F = 1.25 dB is slightly higher than the simulated noise figure (0.87 dB). This on-body gain measurement shows that the active circuitry is not much affected by the human body despite it being directed toward the body without shielding. This is because the amplifier does not radiate and has little interaction with the human body. Figure 4.7 shows the simulated and measured stability factors K and B₁. In general, good agreement is observed, with the LNA meeting all the conditions for unconditional stability (K>1 and B₁ > 0).

Figure 4.6 Simulated and measured amplifier gain and noise figure for the LNA in Fig. 4.3. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

Figure 4.7 Simulated and measured stability factors K and B₁ for the LNA in Fig. 4.3. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

Off-Body Active Antenna Performance

Amplifier Gain and Noise Figure Measurements of an Amplifying Antenna

Now, the input port of the amplifier is connected to the optimized antenna. As the integrated antenna is now a one-port device, the conventional two-port method can no longer be used to measure the amplifier gain and noise figure. In 1994, An et al. (22) proposed a simple measurement method for receiving amplifying antennas. This method is adopted here. Figure 4.8 shows the configuration of a receiving active amplifying antenna. The transducer gain (G_T) is defined as

4.5

where P_av is the available power received by the antenna (with input impedance Z_ant) and P_L is the power delivered to the load (with input impedance Z_L). The total available gain (G_A) of the active antenna can then be calculated:

4.6 G_p, the antenna gain of the passive antenna, can be defined as

4.7

where A_e(θ, ϕ) is the effective aperture of the passive antenna.

Figure 4.8 Configuration of a receiving amplifying antenna.

The measurement is conducted inside an anechoic chamber, with the setup shown in Fig. 4.9. A transmitting standard gain horn is connected to port 1 of a VNA, and the receiving active amplifying antenna is connected to port 2. A similar passive antenna is also fabricated by removing the amplifier part. The distance (L) between the horn and the antenna under test must satisfy the far-field condition (L > 10λ and L > 2D²/λ), where D is the maximum dimension of the antenna and λ is the operating wavelength. The transmission coefficients of the passive and active antennas are then measured, along with the reflection coefficient of the passive antenna. The Friis transmission equation states that S_21a transmitting from the horn to the active antenna can be expressed as

4.8

where G_t is the antenna gain of of the standard gain horn.

If the active antenna is replaced by the passive antenna,

4.9

where S_11p is the reflection coefficient of the passive antenna. Subtracting (4.9) from (4.8), we obtain

4.10

Therefore,

4.11

The transducer gain G_T can easily be calculated from (4.11) by knowing the reflection and transmission coefficients of the active and passive antennas.

Figure 4.9 Measurement setup for amplifying antenna [22].

Next, the noise figure measurement is discussed. With the knowledge of G_T, the noise factor of the receiving active amplifying antenna can be calculated using (4.12). The noise figure is defined as NF = log₁₀F.

4.12

where P_n is the absolute power of noise density that can be measured directly by a noise figure meter, and T_a is the ambient room temperature.

Available Gain and Noise Figure

In the measurement setup (Fig. 4.9), an Agilent N5242A PNA-X vector network analyzer (VNA) was used to measure the transmission (S_21a, S_21p) and reflection (S_11p) coefficients in an anechoic chamber. The absolute noise power density P_n was measured using the calibrated receiver of the PNA-X VNA. Measuring P_n at the output of the active antenna requires a long cable which has been included in the calibration. Inside the anechoic chamber, T_a is usually taken to be 300 K. The simulated and measured results are compared in Fig. 4.10, with good agreement. Using (4.6) and (4.11), the measured available gain G_A of the receiving active amplifying antenna is about 12 dB in the ISM frequency band (2.4 to 2.484 GHz), with a ripple amplitude of 0.5 dB. The simulated and measured noise figures are also given in Fig. 4.10 in the frequency range 2 to 3 GHz. An NF of about 1.3 dB is measured in the ISM band.

Figure 4.10 Simulated and measured NF and G_A of the active amplifying antenna in Fig. 4.5. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

On-Body Active Antenna Performance

To further understand the effect of a human body on an antenna, an active antenna is placed inside a firefighter's jacket made of the same fabric materials. The antenna is placed at two positions (1 and 2) on the person's back (19). For comparison, a similar passive antenna is measured at the same positions. Measurements were conducted in an anechoic chamber using the experimental setup described in Fig. 4.9. The free space and on-body total gains G_tot (or the total available gain G_A) of the active antenna were measured in the frequency range 2 to 3 GHz at both positions on the back, with the results shown in Fig. 4.11. Also depicted in the same figure are the antenna gains of the passive antenna at the two positions. The maximum free-space G_tot of the receiving amplifying antenna is about 17 dBi. As can be seen from the figure, human body has caused 1 to 3 dB of power loss to G_tot, depending on the position at which the antenna is placed on the body. It is also noted that the human body causes a noticeable degradation (about 1 to 3 dB) of the antenna gain. It is worth mentioning that position 1 has a higher loss because it is placed closer to the skin. Figure 4.12 shows the simulated and measured reflection coefficients in free space and on the body. As can be seen from the figure, the human body does not greatly affect the impedance matching. The discrepancy between the simulated and measured results can be caused by various experimental tolerances, which are quite significant in this case.

Figure 4.11 Measured gains of a receiving amplifying antenna at different body positions. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

Figure 4.12 Simulated and measured reflection coefficients of the receiving amplifying antenna in Fig. 4.5 at various settings. (Courtesy of F. Declercq, Ghent University. From (19), copyright © 2010 IEEE, with permission.)

4.4 Active Broadband Low-Noise Amplifying Antenna

Configuration

In this section, the resistive equalization technique is used for co-designing and co-optimizing a broadband amplifying patch antenna. With the use of two equalizing resistors, the input impedance of a dual-band patch antenna can be configured to locate within the region where the field-effect transistor (FET) has the optimum noise figure. To demonstrate the design idea, a broadband receiving antenna is designed to cover the DCS-1800 and UMTS frequency bands.

Antenna Design

The double-layered patch antenna reported by Rajo-Iglesias et al. (24) will be adopted for designing the following amplifying antenna. Figure 4.13 shows the antenna configuration, along with the design parameters: W = 38 mm, h = 6 mm, d₁ = 10 mm, h = 6 mm, and ε _r = 3. It has a ground size of 260 × 260 mm². Given in Fig. 4.14(a) is the measured input impedance of the passive antenna with d₂ = 4 mm. As can be seen clearly from the figure, the antenna has two resonances, at 1.91 and 2.192 GHz. The antenna gain is about 7 ± 1.5 dB. Also shown on the same figure is the impedance region in which the FET has optimum noise (discussed in the next section). With reference to Fig. 4.14(a), unfortunately, the impedance locus of the antenna is far away from the optimum noise impedance region of the FET, which is very undesirable. It causes the amplifier to have a higher noise figure. By increasing the offset of the top–bottom patches to d₂ = 9 mm, shown in Fig. 4.14(b), a small portion of the impedance locus (around the resonances) can be moved into the optimum region. It will later be shown in the co-design process that the use of equalizing resistors can squeeze even more of the impedance locus of the antenna into the optimum noise impedance region.

Figure 4.13 Double-layered wideband patch antenna (24).

Figure 4.14 Input impedance of the double-layered antenna in Fig. 4.13 with (a) d₂ = 4 mm; (b) d₂ = 9 mm. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Amplifier Design

The ATF34143 MESFET (Avagotech, San Jose, CA) is deployed for designing a low-noise amplifier. At a dc bias of V_ds = 3 V and I_ds = 20 mA (25), the transistor has a minimum noise figure of F_min = 0.17 dB at 1.8 GHz, with the input port optimized to have Γ_s = Γ_opt = 074∠57°. When applying the resistive equalization technique, the antenna is made to provide this impedance Γ_ant = Γ_opt. The small-signal model (inside the dashed square) of the FET is shown in Fig. 4.15, along with all the parasitics. As can be seen from the figure, g_m is the transconductance of the transistor. R_g = R_gate and R_d = R_drain are the external resistors that will be used for resistive equalization later.

Figure 4.15 Equivalent circuit of an FET at low frequencies. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Co-design of the Active Broadband Receiving Antenna

By connecting the antenna to the gate of the FET, a broadband amplifying antenna can be designed for the receiver, as shown in Fig. 4.16. The biasing voltages and current of the FET are V_ds = 3 V, V_gs = − 0.5 V, and I_ds = 20 mA. The prototype (circuit side) is shown in Fig. 4.17. The antenna is built on the opposite side (not shown). It is obvious that no input-matching network is used for this active circuit. With reference to the figure, two resistors, R_g and R_d, have been added to the amplifier to optimize the antenna efficiency and noise figure. Now, the resistive equalization technique is applied to meet the following design criteria:

Figure 4.16 Configuration of a broadband amplifying antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Figure 4.17 Broadband amplifying antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

As the input impedance of the antenna is complex and far from the resonance conditions, the generalized S parameters (26, 27) can be used to analyze the network, shown in Fig. 4.18. Assuming that the input and output ports of the amplifier are ports 1 and 2, respectively, the matching condition at port 1 (Z_ant = Z_in^*) implies that

4.13

where [S] denotes the S-parameter matrix of the FET.

Figure 4.18 Generalized S parameters for a broadband amplifying antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

By assuming that Z_L = Z₀ and Γ_L = 0 (Z₀ is the reference characteristic impedance), the generalized S parameters of the network in Fig. 4.18 can be written as

4.14

Conditions (4.13) and (4.14) imply that . Also, it is equal to the optimum noise impedance of the transistor, Γ_ant = Γ_opt. Considering the lossy small-signal model in Fig. 4.15, the inclusion of lossy shunt elements to the gate (Y_gate) and drain (Y_drain) of the FET leads to the new S₁₁ expression.

4.15

where Y_ij are the Y parameters of the FET. For the common case G_drain|Y₂₁|²R_n, the minimum noise figure is given by

4.16

where R_n is the resistance of the equivalent noise voltage generator and G_corr is the correlated admittance between the voltage and current noise generators. The total noise figure can then be expressed as

4.17

where G_opt and B_opt are defined as

4.18

It can be seen from eq. (4.18) that the noise figure is affected by the gate conductance. The effect of gate conductance is studied in Fig. 4.19. As can be seen from the figure, a higher R_g/Z₀ value is good for a lower noise figure.

Figure 4.19 Effect of the gate conductance on F^total across the frequency. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Using the model in Fig. 4.18, the transducer gain (G_T) can be defined as

4.19

4.20

where the transmission coefficient S₂₁ is defined as

4.21

4.4.1.2 Results and Discussion

The AWR Microwave Office was used to optimize the models in Figs. 4.15 and 4.16, and it was found that the optimized drain and gate resistances are R_d = R_drain = 91 Ω and R_g = R_gate = 68 Ω, respectively. With reference to Fig. 4.16, two high-impedance lines (Z₀ = 90 Ω, with an electrical length of 90°) are used to compensate the effect of losses at higher frequencies. The new input impedance of the antenna (inclusive of R_d and R_g) is shown in Fig. 4.20. The addition of the two resistors causes the locus of the antenna impedance to become narrower. Therefore, a larger portion of the locus can then be squeezed into the region in which the FET has optimum noise. Also plotted on the figure is the locus of the optimum noise impedance of the FET. It can be seen from the figure that better impedance matching can now be obtained as the two loci are closer to each other [compared to Fig. 4.14(b)]. These two resistors are called equalizing resistors.

Figure 4.20 New input impedance of a double-layered patch antenna (with inclusion of R_d and R_g). Also given is the locus of the optimum noise impedance of the FET. Z. Trans refers to the optimum noise impedance of the transistor and Z. Patch refers to the new input impedance of the antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Now, the gain of the active antenna is characterized. For an active amplifying antenna, its effective transmission gain (G_TX) can be obtained by comparing its transmission coefficient to that of its passive counterpart. Of course, G_TX is a figure that combines the antenna and amplifier gains. A standard reference horn is used for measurement in an anechoic chamber. With reference to Fig. 4.21, the measured G_TX( = ΔS₂₁) reads about 13 dB across the frequency passband. It is then compared with the simulated G_T of the isolated amplifier in Fig. 4.22. It is worth mentioning that the antenna impedance is used as the port impedance at the input. As can be seen from the figure, the simulated G_T agrees reasonably well with the measured G_TX.

Figure 4.21 Measured effective transmission gain (G_TX) of active and passive antennas. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Figure 4.22 Comparison of the simulated transducer gain (G_T) and the measured effective transmission gain (G_TX). (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

The ratio of antenna gain to effective system temperature (G/T) ratio is a figure of merit that describes the performance of an AIA (28, 29). For an active antenna, its G and T cannot be measured separately. The G/T ratio is defined as

4.22

with the following parameters:

Once G/T is known, the total noise figure of the active antenna can be calculated:

4.23

Figure 4.23(a) shows the measurement setup for the active antenna in an anechoic chamber. With reference to the figure, a section of 50-Ω transmission line (with a loss of 2 dB) is used to connect the active antenna to the receiver (with a loss of 8 dB). The passive antenna can be measured by the setup in Fig. 4.23(b). In this case, the antenna is connected to an external LNA through a section of transmission line (with a loss of 2 dB). Figure 4.24 shows the measured G/T of the active antenna for different (V_ds, V_gs) biasing conditions. Several combinations (1.5V, 0.5V), (2.5V, 0.5V), (3.0V, 0.6V), and (3.5V, 0.6V) were tested. Also shown on the figure is the measured G/T of the passive antenna. When biased at V_ds = 2.5 V and V_gs = 0.5 V, the ratio G/T is about −16 dB/K in the frequency range 1.5 to 2.05 GHz (bandwidth of ∼30%), as can be seen from Fig. 4.24. It is 4 to 6 dB better than that for the passive antenna.

Figure 4.23 Measurement setups for (a) an active antenna; (b) a passive antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Figure 4.24 Measured G/T of the active antenna in Fig. 4.16 with different biasing conditions (V_ds and V_gs). Also shown is the measured G/T of a passive antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

By letting the antenna temperature T₀ = T_a = 290 K and knowing the G/T ratio, the noise figure NF can be calculated using (4.23). Figure 4.25 shows the NF calculated from the simulated and measured G/T values. With reference to the figure, the NF measured at 1.8 GHz is about 0.5 dB, being higher than that given in the datasheet (0.17 dB).

Figure 4.25 Comparison of noise figures (NFs) calculated from simulated and measured G/T. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Figure 4.26 shows the simulated and measured reflection coefficients of the broadband amplifying antenna. Since port 1 of the amplifier is now connected to the antenna, the S₂₂ notation is used to denote the reflection coefficient at the new input port of the active antenna. Three resonances are observed in both the simulation and measurement, with reasonable agreement. Also shown in the figure is the measured reflection coefficient of the passive antenna. It is higher because the antenna works in the nonresonant modes. Figure 4.27 shows the radiation patterns measured for the active and passive antennas at 1.8 and 2.2 GHz, both in the E-plane. They have almost the same field patterns, implying that the amplifier does not greatly disturb the far-field characteristics of the passive antenna.

Figure 4.26 Simulated and measured reflection coefficients (S₂₂) of the active antenna in Fig. 4.16. Also shown is the S₂₂ measured for a passive antenna. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

Figure 4.27 Co-polarized fields measured for active (solid line) and passive (dashed line) antennas in the E-plane. (Courtesy of D. Segovia-Vargas, Carlos III University. From (23), copyright © 2008 IEEE, with permission.)

4.5 Conclusions

In this chapter, the co-design and co-optimization processes of receiving amplifying antennas have been explored. In the first part, the amplifying antenna is made on textile and is wearable. The interaction between an antenna, an LNA, and the human body has been studied. It has been found in the second part that wide impedance matching is achievable by deploying two equalization resistors to the gate and drain of a FET. For both parts, the antenna performance is not greatly affected by the amplifier.

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