Chapter 13

Microphone Preamplifiers

Microphone Preamplifier Requirements

A microphone preamplifier is a serious design challenge. It must provide a gain variable from 0 to 70 dB or more to amplify deafening drum-kits or discrete dulcimers, generate minimal internal noise, and have a high common-mode rejection ratio (CMRR) balanced input to cancel noise pickup in long cables. It must also be able to withstand +48 V DC of phantom power suddenly applied to its input while handling microvolt signals.

It is now rare to use input transformers to match a low-impedance (150–200 Ω) microphone to the preamplifier, since the cost and weight penalty is serious, especially when linearity at low frequencies and high levels is important. Both dynamic and capacitor microphones have a low output impedance of this order, the former because of the low number of turns used in the coil, the latter because active buffering of the extremely high capsule impedance is essential.

The low-noise requirement rules out the direct use of op-amps, since their design involves compromises that make them at least 10 dB noisier than discrete transistors when faced with low source impedances. The answer, for at least the last 30 years, has been to use hybrid input stages that combine discrete input transistors to give low noise, combined with op-amps to provide raw open-loop gain for linearization, and load-driving capability.

The requirements of a microphone preamplifier can be summarized as follows:

1. Variable gain is necessary, usually from 0 to +70 dB. Some designs have a gain range extending to +80 dB. The bottom 20 dB section of the range is often accessed by switching in a 20 dB input attenuator.

2. Minimal noise must be produced. Taking the source impedance of a microphone as 200 Ω, the Johnson noise from that resistance is −129.6 dBu at 25^ºC, 20 kHz bandwidth. This puts an immediate limit on the noise performance; if you set up 70 dB of gain then the noise output from the preamplifier will be −59.6 dBu even if it is itself completely noiseless. Most mic preamplifiers approach this at maximum gain, often having a noise figure as low as 1 or 2 dB, but depart from it further and further as gain is reduced. In other words, the noise output does not fall as fast as it would if the preamp was noiseless, because reducing the gain means increasing the effective resistance of the gain control network, so it creates more Johnson noise.

3. The input must have a high CMRR to reject interference and ground noise. The CMRR should ideally be high, be flat with frequency, and remain high as the input gain is altered over the whole range. In practice, CMRR tends to worsen as gain is reduced. Because of the need for a balanced input, microphone inputs are almost always female XLR connectors.

4. The input must have a constant resistive input impedance of 1−2 kΩ, which provides appropriate loading for a 200 Ω dynamic microphone capsule. This is also a suitable load for the internal head amplifiers of capacitor microphones.

5. The input must be proofed against the sudden application (or removal) of +48 V DC phantom power. It must withstand this for many repeated cycles over the life of the equipment.

Transformer Microphone Inputs

For a long time transformer microphone inputs were the only real option. The cost and weight of a microphone input transformer for every channel are considerable, especially at the quality end of the market, because a large transformer core is needed to handle high levels at low frequency without distortion. Because of the low signal levels, mu-metal screening cans were normally used to minimize magnetic interference, and this added to the cost and weight.

Step-up ratios from 1:5 to 1:10 were used, the higher ratios giving a better noise performance but more difficulties with frequency response because of the greater capacitance of a larger secondary winding. The impedance reflected to the input depends on the square of the step-up ratio, so with a 1:10 transformer the secondary had to be loaded with 100 kΩ to get the desired 1 kΩ at the primary input.

The arrangement in Figure 13.1 has an amplifier gain range from 0.3 to 60 dB, to which is added 20 dB from the transformer ratio of 1:10, giving +20 to +80 dB of overall gain. R6 in conjunction with the reverse-log pot shapes the control law so it is roughly logarithmic. R4 is the 100 kΩ secondary loading resistor, and C1, R3 form a Zobel network to damp the secondary resonance. In the mid-1970s, when op-amps were still a dubious proposition for quality audio, the input amplifiers were very often discrete transistor stages using four or more transistors. Note R5, which prevents the gain from being reduced to exactly unity. Its presence is testimony to the fact that the discrete preamplifiers were difficult to stabilize at HF if the output and non-inverting input were directly connected together. R1 and R2 feed in phantom power when the switch is in the On position.

Microphone transformer technology had, and still has, its own advantages. A balanced input with a good and constant CMRR was inherent. The step-up ratio, which gave ‘gain for free’ in electronic if not financial terms, and the associated impedance matching, meant that it was easy to design a quiet input stage. Transformers also gave good RF rejection, and isolated the input preamplifier from phantom power voltages. For EMC reasons in particular, transformer microphones continued in use in broadcast mixing consoles long after they had disappeared from recording and PA equipment.

The Simple Hybrid Microphone Preamplifier

The cost incentive to develop an effective low-noise transformerless microphone input was considerable, and after much experimentation the arrangement shown here in Figure 13.2 became pretty much standard for some years, being extensively used in mixers around the period 1978–1984 (in this period the more expensive consoles stuck to transformer microphone amplifiers). The difficulties of getting the noise low enough and the linearity good enough were at first formidable, but after a good deal of work (some of which I did) such stages have become almost universal. Q1 and Q2 work as common-emitter stages, with the gain-control network R5, RV1, C3 connected between the emitters. As the resistance of this network is reduced, the differential gain increases but the common-mode gain remains low. The two signals at the collectors are then summed by the op-amp. This does not have to be a low-noise type because the gain in the transistors means it is they which determine the noise performance, and the TL072 was almost universally used in this position, being at the time much cheaper than the 5532. With appropriate choice of transistors (a type with low R_b) and collector current, preamplifiers of this type can give an equivalent input noise (EIN) of −128 dBu at maximum gain. The EIN rises as gain is reduced, because the resistance of the gain-control network and its Johnson noise is increased. The noise output, however, still falls as gain is reduced.

The gain law is very non-linear with R_g; a reverse-log D-law pot helps but there is still some cramping of the calibration at the high gain end, and preamplifier gain ranges of more than 50 dB are not really practicable with this configuration.

Today this approach is considered obsolete for anything except budget mixer purposes because of the mediocre distortion performance. You will note that the input transistor pair have no overall feedback loop closed around them, and their non-linearity creates significant distortion, especially at high gains.

The two reverse-biased diodes in the transistor collector circuits are to prevent the op-amp being damaged by having its input driven below the −17 V rail when phantom power is applied or removed. Note that the input coupling capacitors C1, C2 shown are not intended to cope with the phantom voltage if used. On arriving at the input XLR, the microphone signal first encounters the phantom feed resistors, DC-blocking capacitors (usually rated at 63 V), a switchable 20 dB attenuator, and possibly a phase-invert switch that swaps over the inputs, before it reaches the preamplifier. This is illustrated below in Figure 13.4.

The Balanced-Feedback Hybrid Microphone Preamplifier

The microphone preamp architecture described in the previous section has the merit of simplicity, but because there is no global feedback loop around the input transistors, its distortion performance falls far short of the rest of a mixer, which typically consists of pure op-amp circuitry with very low distortion. It is obviously undesirable practically, aesthetically, and in every other way for the very first stage in the signal chain to irretrievably mess up the signal, but this is what happened for several years. In the mid-1980s I decided to do something about this. Many methods of applying feedback to the input stage were tried, but foundered on the fact that if feedback was applied to one of the transistors, the current variations in the other were excessive and created distortion. The use of two feedback paths in anti-phase, i.e. balanced feedback, was the solution to this problem, but this meant that one feedback path would have to be via an inverting amplifier with extra phase-shifts that would imperil HF stability.

The basic concept is shown in Figure 13.3. Direct negative feedback to Q2 goes through R12, while the inverted feedback passes through the unity-gain inverting stage U1:B and goes through R11 to Q1. The gain-control network R5, RV1, and C3 is connected between the two feedback points at the Q1, Q2 emitter, and sets the closed-loop gain by controlling the NFB factor. Distortion is pretty much eliminated, even at maximum gain.

The solution to the stability problem is to make sure that the direct HF feedback through C10 dominates that through C12, which has been through the inverter U1:A; this is aided by adding C4 to the inverter to control its HF response. This feedback is therefore not symmetrical at HF, but this has no effect on the functioning of the circuit with audio signals.

The closed-loop gain is given by Equation 13.1, which looks complicated, but becomes clearer when you appreciate that the terms in brackets simply represent the parallel combination of the emitter resistor R_e (R1, R2 in the figure) and half the resistance of the gain control network R_g. R_NFB is the value of R11, R12. The 2 on the bottom comes from the fact that we are only using one output of the amplifier – there is an inverted output from the inverter, which could be phase-summed to give twice the gain. We don’t do that because it is actually desirable to have as low a minimum gain as possible. With the values shown the gain range is +22 to +71 dB, which in conjunction with a switchable 20 dB pad gives a useful total range of about 0−70 dB.

In order to get enough maximum gain, the feedback resistors R11, R12 have to be quite high at 82 kΩ. This means that the control of the DC level at the output is not very good. To solve this the DC servo integrator U2:A was added; R13 is connected to the preamplifier output and the integrator acts via the non-inverting input of the inverter op-amp to keep the final output at 0 V.

This technology was used on several consoles, such as the Soundcraft TS12, but it had a relatively short life as I came up with something better – the padless microphone preamplifier described later in this chapter.

Microphone and Line-Input Pads

Microphone pads, or attenuators, are used when the output is too high for the mixing console input to cope with; this typically happens when you put a microphone inside a kick-drum. Attenuators are also used when for reasons of economy it is desirable that the microphone input doubles up as a line input. A typical arrangement is shown in Figure 13.4. The preamplifier has a typical gain range of + 20 to +70 dB. There is an input XLR and phantom feed resistors R1, R2. C1 and C2 are DC-blocking capacitors to stop the phantom voltage from getting into the preamplifier; these should be as large as possible to preserve LF CMRR. Next comes a 20 dB balanced attenuator made up of R3–R6; note that the loading of the preamplifier input resistor R10 must be taken into account when designing the attenuator resistor values; one of the functions of this resistor is to prevent the preamplifier input from being open-circuited when the pad switch SW1 or the mic/line switch SW2 is between contacts. C3 and C4 are two further DC-blocking capacitors that prevent the input terminals of the preamplifier, which are not in general at 0 V, from causing clicks in the switching. C5, C6, and C7 increase EMC immunity and also keep the preamplifier from oscillating if the microphone input is left open-circuit at maximum gain. Such oscillation is not an indication that the preamplifier itself is unstable – it normally happens because the insert jacks, which carry the output signal from the preamplifier, are capacitatively crosstalking to the microphone input, forming a feedback loop. Ideally the microphone input should be open-circuit stable not only with the input gain at maximum, but also with full treble boost set up on the EQ section. This can be challenging to achieve, but it is possible. I have done it many times.

In line mode, the microphone gain is of course much too high, and the usual practice is to use a 30 dB attenuator on the line input, which allows a high input impedance to be set by R7, R9, while R8 provides a low source impedance to minimize preamplifier noise. This pad alters the gain range to −10 to +40 dB, which is actually too wide for a line input, and some consoles have another switch section in the mic/line switch, which reduces the gain range of the preamplifier so that the overall range is a more useful −10 to +20 dB. The larger and more expensive consoles usually have separate line-input stages, which avoid the compromises inherent in using the microphone input as a line input.

There is an important point to be made about the two attenuators. You will have noticed that the microphone attenuator uses four resistors and has its center connected to ground, whereas the line attenuator uses a more economical three resistors with no ground connection.

The disadvantage of the three-resistor version is that the wanted differential signal is attenuated, but the unwanted common-mode signal is not, and so the CMRR is much worsened. This does not happen with the four-resistor configuration because the ground connection means that both differential and common-mode signals are attenuated equally. There is no reason why the line attenuator here could not have been designed with four resistors – I just wanted to make the point.

The microphone amplifiers described have a high CMRR, and a problem with attenuators like this is that both types degrade the overall CMRR quite seriously because of their resistor tolerances, even if 1% components are used.

The Padless Microphone Preamplifier

The ideal microphone preamplifier would have a gain range of 70 dB or thereabouts on a single control, going down to unity gain without the inconvenience of a pad switch. It was mentioned in the previous section that resistive pads degrade the overall CMRR, and also the noise performance, as an inevitable consequence of following a 20 dB pad with an amplifier having 20 dB of gain. In addition, space on a channel front panel is always in short supply and losing a switch would be very welcome. I therefore invented the padless microphone preamplifier. Looking at the mixer market today (2009), the idea seems to have caught on.

The concept is based on the balanced-feedback mic amp described above, but now the total gain is spread over two stages to give a smooth 0−70 dB gain range with the rotation of a single knob.

The first stage shown in Figure 13.5 is based on the BFMA circuit in Figure 13.3, but with the feedback resistors reduced to 2k7 to reduce the gain range. The gain-control network R5, RV1, and C3 has also been halved in resistance to reduce Johnson noise, and the net result is a gain range of +1.5 to +49 dB; as before a reverse-log D-law pot is used. The lower feedback resistors mean that no servo is required to correct the DC conditions. Note that the greater amount of NFB means that under overload conditions it is possible for the common-mode range of the op-amp to be exceeded, leading to the well-known TL072 phase reversal and latch-up. This is prevented by R11, D3, and D4, which have no effect on linearity in normal operation.

The second stage of the padless mic amp is shown in Figure 13.6. This consists essentially of a variable-gain balanced input stage as described in Chapter 14, configured for a gain range of 0 to +20 dB. The gain pot is once again a reverse-log D-law pot and the combination of the gain laws of the two stages gives a very reasonable law over the almost 70 dB range, though there is still a little cramping at the high-gain end.

The second stage is also used as a line-input stage with a gain range of −10 to +20 dB. The mic/line switching used to do this may look rather complex but it does a bit more than simply change sources. In Figure 13.6 switch SW1 is shown in the ‘mic’ position, and the first stage reaches the inverting input of the second via SW1-B; the output of the mic amp is phase-inverted simply by swapping over its inputs. Line-input resistors R1, R2 give reduced gain for line input working, and in mic mode they are shorted together by SW1-A to prevent crosstalk from line to mic, which is an important issue when track normalling is incorporated (see Chapter 12). In several of my designs these resistors were placed on the input connector PCB rather than in the channel, to keep their hot ends away from other circuitry and further reduce line to mic crosstalk. SW1-C shorts the junction of R1, R2 to ground, further improving mic/ line crosstalk if the line input is not balanced. SW1-D shorts the unused second-stage noninverting input to ground.

In line mode R1, R2 are connected to the second stage via SW1-B and SW1-C. In mic mode, SW1-D shorts R8 to ground so that the gain range of the second stage is increased to the required 30 dB. SW2 is a phase-invert switch that simply swaps over the input connections to the second stage.

The padless mic amp gives both a good law gain control and lower noise at low gain settings. The noise performance versus gain for a typical example can be seen in Table 13.1; as described earlier, the EIN and noise figures worsen as the gain is turned down, due to the increased resistance of the gain control network. A noise figure of 30 dB may appear to be pretty dire, but the corresponding noise output is only −98 dBu, and this will soon be submerged in the noise from the following stages.

This effect can be reduced by reducing the impedance of the feedback and gain-control network, but this increases the power required to drive them, and because of the square root in the Johnson noise equation, a reduction by a factor of 10, which would need some serious electronics, would only give a 10 dB improvement, and that at the low gain end where it is least needed. Specialized outboard mic amps with low-resistance feedback networks driven by what are in effect small power amplifiers have been developed but do not seem to have caught on.

Another advantage of the padless approach is that one pair of DC-blocking capacitors suffices, rated at 63 V as in Figure 13.5, and this improves the CMRR at low frequencies. The padless microphone preamplifier concept was protected by patent number GB 2242089 in 1991, and was used extensively over many ranges of mixing console.

Capacitor Microphone Head Amplifiers

A capacitor capsule has an extremely high output impedance, equivalent to a very small capacitor of a few picofarads. It is in fact the highest impedance you are ever likely to encounter in the audio business, and certainly the highest I have ever had to deal with. Special circuit techniques are required to combine low noise and high impedance, working with a strictly limited amount of power. A while ago I designed the electronics for a new capacitor microphone by one of the well-known manufacturers, and the circuitry described here is a somewhat simplified version of that.

The first point is that the microphone capsule had an impedance of about 5 pF, so to get a −3 dB point of 10 Hz the total load impedance has to be no more than 3.2 GΩ (yes, that’s 3200 MΩ). The capsule needs to be fed with a polarizing voltage through a resistor, and the head amplifier needs a biasing resistor. In this design both were 10 GΩ, which means that the input impedance of the amplifier itself had to be not less than 8.9 GΩ. Resistors with these astronomical values are exotic components that come in a glass encapsulation that must be manipulated with tweezers – one touch of a finger and the insulation properties of the glass are fatally compromised.

Figure 13.7 shows my capacitor mic head amp. R1 supplies the capsule polarizing voltage and R2 biases the first stage, a unity-gain JFET source-follower augmented by op-amp U1:A, which provides the gain for a high NFB factor to linearize Q1. The drain of Q1 is bootstrapped via C3 to prevent local feedback through the gate-drain capacitance of the JFET from reducing the input impedance. R3, R5, R6 set the DC conditions for Q1.

The second stage is a low-noise amplifier with gain of +4 dB, defined by the ratio of feedback resistor R17 to R15 and R16. Like the first stage, it is a hybrid design that combines the low noise of low-R_b transistor Q2 with the open-loop gain and load-driving capability of an op-amp.

The stage also acts as a unity-gain follower, making up a second-order Butterworth Sallenand-Key high-pass filter when C6, C7 or C8, C9 are switched in; the resistive elements are R10 and R14.

The system has two steps of attenuation: −10 and −20 dB. The first 10 dB step is obtained by using SW1-B to take the output from the junction of R15 and R16 instead of the normal stage output. The second 10 dB step results from switching C2 across the mic capsule, forming a capacitative attenuator that reduces the input to the first stage and prevents overload. The gains available are thus +4, −6, and −16 dB. The maximum sound pressure handling is +146 dB SPL, or +155 dB SPL with the −20 dB pad engaged.

The incoming phantom power is tapped off by R20, R21 and fed to a discrete BJT regulator that gives +32 V to power the op-amp, and +14 V to run a small LC oscillator that pumps the + 32 V rail up to the +63 V required to polarize the capsule. Total current consumption is 2.2 mA.

The noise output is −120.7 dBu (A-weighted), which may appear high compared with the microphone amplifiers described above, but remember that a capacitor microphone puts out a high signal voltage so that the signal-to-noise ratio is actually very good. The mic capsule, being a pure reactance, generates no noise of its own, and its noise output comes only from the Brownian motion of the air against the capsule diaphragm and from the electronics. Noise measurements of this technology require special methods. The impedances are so high that meaningful results can only be obtained by putting the circuitry inside a completely closed metal screening enclosure.