2.4.1 Methods for Generating Pulse Trains and Transmitter Sources

In the 1970s, Harmuth (1972a, pp. 244-291, 1977b, pp. 235-399) discussed a variety of impulse radiators and presented approaches to practical radiators. Examples of radiators and selective receivers were discussed at this time.

2.4.2 Methods for Modulating a Pulse Train

A variety of methods for modulating a pulse train have been known for decades, even before the age of transistors. Harmuth (1969, 1972a, 1972b, 1972c) and (Smith, 1966, p. 438) have addressed some of the early methods.

Some of the many methods available for transmitter sources include the following:

• Light-activated semiconductor switches (LASS)—these switches are generally Si-based (cf. Auston, 1975; Loubriel et al., 1993; Mourou and Knox, 1979; Nunnally and Edwards, 1991; Kingsley et al., 1995).

• Light-activated bulk avalanche semiconductor switches (BASS)—these switches are generally GaAs-based (cf. Jayarman and Lee, 1972; Kingsley et al., 1995; Loubriel et al., 1993, 1995; Pocha et al., 1991; Sarkar et al. 1993; Vainshtein et al., 1988).

• Thyristors based on GaAs (cf. Platts et al., 1995), semiconductor-based pulse compressor systems (cf. Edwards et al., 1995), and Marx bank pulse generators (cf. Edwards et al., 1995; Platts, 1991; Platts et al., 1995).

• Avalanche drift diode generators (cf. Edwards et al., 1995; Grekhov et al., 1981, 1985).

• Vacuum triodes (cf. Platts et al., 1995).

• Magnetic switches (cf. Platts et al., 1995).

• Low-voltage tunnel diodes (cf. Ross, 1973a, 1973b; Ross and Lamensdorf, 1972; Ross and Robbins, 1973).

• High-voltage avalanche semiconductor diodes, laser diodes, and resonant microwave compressors (Didenko and Novikov, 1991; Yushkov and Badulin, 1997).

• Avalanche transistor/diode methods of switching: A simple method of generating UWB signals is to use avalanche transistors, and it has been known for many years. Morey (1974) cited a transistor in the avalanche mode as a suitable means for a pulse generator in a UWB system.

• Table 2.1 from Ross (1986) lists the avalanche transistor as one method (among others) for achieving pulse sources. Andrews (1986) reviewed the field of fast pulse generators and, in particular, Picosecond Pulse Labs avalanche transistor pulse generators (p. 103). Astanin and Kostylev (1989) state that “generators based on avalanche transistors are widely used” (p. 108), thereby making clear that the usage of such generators was not restricted to the former Soviet Union.

  TABLE 2.1
  Typical Characteristics of State-of-the-Art Pulse Sources in 1986

  

• One method involves light-activated processes for switching generators—Silicon-made RF short-pulse generators using laser-induced photoconductivity in high-resistivity semiconductors were demonstrated by Auston (1975). Linear photoconductive semiconductive switching has been an active field since that time (cf. Lee, 1984; Rosen and Zutavern, 1994). A variety of semiconductors have been employed in photoconductive switches, for example, Si, GaAs, ZnSe, diamond, and SiC (cf. Kingsley et al., 1995). The activity in the field of light-activated switching has increased to such an extent that almost every year meetings on this topic are held by the SPIE. The patent by Kim et al. (1993) also addresses light-activated switching for impulse systems.

2.4.3 Methods for Pulse Detection and Reception

Methods for pulse detection and receiving include leading-edge detection, sampling bridge circuit s, monostable multivibrators, integration and averaging, templ ate signal match detection, correlation detection techniques, signal integration, and synchronous detection. All such methods have been known since the 1960s and 1970s [cf. Malmstadt and Enke (1963) for a review of early examples of these methods]. A nonexhaustive list of these well-known methods includes leading-edge detection, which has long been in use (cf. Meleshko, 1987, p.58), and sampling bridge circuit, commonplace by 1968. The Tektronix Instruction Manual for the Type S-2 Sampling Head (1968) provides a sampling bridge circuit. Also included are monostable multivibrator means, a technique that has been widely used for over 50 years (cf. Malmstadt and Enke, 1963, p. 440), integrating and averaging methods—electronic integration and averaging is also a technique that has been commonly used over the last 50 years (cf. Smith, 1966, p. 450)—and the commonplace template signal match detection (e.g., Meleshko, 1987, Figure 32, p. 65). A template match is essentially a logical operation or a cross-correlation satisfying the long-known Wiener-Hopf equation. It should be noted that template pulse mixing of a signal with a gating pulse is not the same as summing a signal with a strobe pulse, and it is also a much less sensitive detection operation.

Correlation detection of RF signals is now universal (cf. Lee, 1960; Skolnik, 1962, p. 275). The sliding correlator as applied to impulse communications and radar was reported in the open literature long ago (cf. Skolnik, 1962, p. 143 and Chapter 6 “Advanced Signal Design and Processing”). Other examples of early sliding correlators were presented by Harmuth (1981, 1984), which are shown in Figure 2.5.

Signal integration methods to match a preset criterion (e.g., summed amplitude) are commonplace. For example, the 1968 Tektronix sampli ng circuit is a transmission gate followed by a short-term integrator (Tektronix, 1968). Synchronous detection methods are widely used and are long-known procedures. Figure 14-86c in the work by Fink and Christiansen (1975, pp. 14-69) shows a product (synchronous) detector. This type of detector has been used since the advent of single-sideband transmission.

In addressing monocycle signals from antennas driven by stepped amplitude (ramp function) driver methods, Cronson (1975) showed that virtually any frequency-coded pulse could be generated by a step function (cf. Ross, 1986, p. 11). This method has also been studied for a long period of time (see Harmuth, 1981, p. 4854, 77; 1990, Preface and “Section 1.7—A Guide to Reading,” p. 52).

Use of subcarriers in pulse trains, that is, subcarrier modulation, is a well-known modulation method and is explained in many textbooks. Transferring the method to the time domain introduces no new principles. The use of subcarriers and subband coding is a well-established field in electrical engineering. Later attempts to patent the well-known modulations such as frequency modulation (FM), amplitude modulation (AM), phase modulation, frequency shift keying (FSK), phase shift keying (PSK), pulse FM, and Manchester coding, if taken seriously, would undermine the field of radio communications inasmuch as these methods are the backbone of this mature field.

2.4.4 Appropriately Efficient UWB Antennas

The antennas used in UWB communications, or being considered for use, include loaded dipoles, transverse electromagnetic (TEM) horns, biconicals, ridged horns, and spiral and large-current antennae. All have a variety of merits and demerits (Allen et al., 2007; Evans and Kong, 1983a, 1983b; Ross, 1968; Shantz, 2005; Smith, 1984; Susman and Lamensdorf, 1971; Theodorou et al., 1981; Wicks and van Etten, 1991; Wohlers, 1970).

2.5.1 The Tektronix™ System (1975)

In 1968, Tektronix offered for sale a sampling head consisting of a strobe generator, a sampling bridge as shown in Figure 2.9, a blow-by and trigger pickoff, and a preamplifier. By 1975, it was possible to build either a UWB communications system or UWB radar using Tektronix laboratory test equipment. Figure 2.10 shows such a system built using a Tektronix 7S12, an S-4 sampler for receiving means, and an AM502 differential amplifier with filters as signal-processing means.

It is instructive to examine the means of achieving detection. The sampling bridge circuit does not amplify, but merely provides, an error signal. Referring to Figure 2.9, the Tektronix manual states as follows:

During the sampling time, the strobe pulses forward bias D5 and D6. By normal bridge function, the conduction of D5 and D6 charges or discharges C5, C6, C7 and C8. The voltage charge on these capacitors changes about 2 1/2 percent of the difference between the feedback and DC offset voltage and the incoming signal voltage. This voltage change, called the error signal, is amplified in the preamplifier.

Thus, diodes D5 and D6 form a sampling bridge or transmission gate driven by strobe pulses at J51 and J53. Resistors R5 and R6 = 200Ω and capacitors C5 and V6 = 5 pF form a short-term integrator (balanced configuration) with a 1-ns time constant. D5 and D6 conduct for approximately 40-50 ps, so that integrators charge to about 2½% of the signal input voltage during the time that the D5 and D6 conduct, providing a voltage-transfer efficiency of 2½%. Therefore, his approach, although much used and imitated, is nonetheless an inefficient detector.

2.5.2 Harmuth Systems

Beginning with publication of the first edition of Transmission of Information by Orthogonal Functions in 1969, Dr. Henning Harmuth addressed UWB in all its manifestations, but under the synonyms impulse, carrier-free, time domain, nonsinusoidal, orthogonal function, Walsh functions, and large-relative-bandwidth radio/radar signals. Remember that the term UWB dates from 1989. The five basic subcomponents were also addressed in early representational form and with the electronics available at that time. The second edition of the same book (Harmuth, 1972a) contains a chapter called “Nonsinusoidal Electromagnetic Waves” (Ch. 6, pp. 244-291), which discussed a variety of impulse radiators and the selective reception of impulse signals in mobile communications (cf. Section 5.4: Signal Selection and Synchronization, pp. 282-291, with a correlation receiver shown in Figure 189 on page 289 [Figure 5.0.1]).

Harmuth (1972a, 1972b, 1972c) described transmitters and selective receivers for periodic waves with arbitrary time variation within the period. Harmuth (1977a) reported a more advanced form of the receiver and discussed “pulse compression”—actually signal averaging. The work of Harmuth (1977b) contains Chapter 3, “Electromagnetic Waves With General Time Variation,” with the sections Practical Radiators, Practical Receivers, and Applications to Radar. Photographs of a radiator and a selective receiver are shown, and oscilloscope recordings of nonsinusoidal waves—specifically their electric field strengths—are also displayed. A correlation receiver for selective reception is shown in Figure 9 of Harmuth (1975).

Harmuth’s Nonsinusoidal Waves for Radar and Radio Communication (1981) contains Chapter 4, “Selective Receivers,” and Section 4.6, Receiver for Nonperiodic Waves. The circuit shown in that book is a correlation circuit. The correlation receiver or pulse compressor was recognized at this time to be the generalized equivalent for waves with arbitrary time variation of the tuned resonant circuit for the selective reception of sinusoidal waves. The pulse compression circuits in this reference have also been published in Harmuth (1979, 1980).

The work of Harmuth (1984) contains Section 1.2, Transmitter and Receiver for Nonsinusoidal Waves; Section 1.3, Nonsinusoidal Spread Spectrum Radio Transmission; and Section 1.4, Pulse Agility Versus Frequency Agility. All of these address the major components of a UWB system.

Thus, by the early 1970s, the generic system and all of the generic subcomponents for UWB systems, of whatever form, were in the public domain, and by the early 1980s, they had been extensively discussed.

2.5.3 Ross and Robbins Systems

Robbins (1972), Robbins and Robbins (1974), Ross and Robbins (1973, 1987), Ross (1973a, 1973b), and Ross and Mara (1994) are all patents addressing specific embodiments of a UWB radio receiver.

The patent by Ross (1973) disclosed an impulse radio encoding intelligence on a train of pulses by pulse interval modulation or pulse position modulation (Ross, 1973, paragraph 9). This patent recognized the utility in communications systems of a wide instantaneous bandwidth (as opposed to the sequential bandwidth). Subsystems of a UWB radio were disclosed by Robbins (1972), Ross and Lamensdorf (1972), and Robbins and Robbins (1974, 1987). Figure 2.11 shows the patented circuit diagram.

These early patents recognized that the pulse train could be modulated by a code scheme and that this method was known even prior to the patents themselves. For example, Ross (1973, paragraph 13) states that

it will be understood by those skilled in the art that a variety of ways is available in the prior art for impressing intelligence on the carrier-less baseband pulses of transmitter 2, and for abstracting that intelligence at receiver 3 by well established demodulation techniques operating on the relatively long pulses generated in receiver 3.

In 1978, Bennett and Ross published Time-Domain Electromagnetics and Its Applications. These authors wrote at a time when the term “baseband” was preferred to “UWB” or “time domain,” but all these terms are synonymous. The essential five major components of a UWB system had been already presented by Ross (1973) using methods of implementation long in use as the following quotation shows:

The baseband pulse receiver 3, as has been seen, may be employed in the novel communication system to receive intelligence communications in a variety of ways, such as . . . Equivalent electronic operation may be readily visualized . . . More sophisticated arrangements for conveying intelligence messages from transmitter 2 to receiver 3 are readily apparent to those skilled in the art.

Ross (1973, paragraph 13, line 20)

In addition:

Similarly, pulse interval modulation in transmitter 2 and cooperative demodulation in receiver 3 may be employed for conveying intelligence messages. It will be understood by those skilled in the art that a variety of ways is available in the prior art for impressing intelligence on the carrier-less base band pulses . . . and for abstracting that intelligence . . . by well established demodulation techniques . . .

Ross (1973, paragraph 13, line 55)

The claims of the Ross (1973) patent refer to particular instances of the five basic components.

2.7.1 Interference Issues

In the case of UWB communications systems, what is new is the assumption that such systems can coexist without interference with other communications systems, which use synchronous receivers, and are regulated by conventional FCC spectrum habitation requirements. This assumption specifically requires that the receivers of conventional systems not only normally operate at higher average power/frequency thresholds than do those of UWB receivers (which must achieve acceptable signal-to-noise levels over time either by signal averaging or by high instantaneous signal power levels) but also are not normally subject to electronic upset by high-peak power, that is, transient UWB signals. These are two separate requirements that are usually assumed identical . Note that the FCC Section 45 regulations state that

. . . UWB devices may not be employed for the operation of toys. Operation onboard an aircraft, a ship or a satellite is prohibited.

This assumed absence of interference of UWB communications systems with other conventional receivers, and also of the electronic upset of a variety of forms of electronic equipment (e.g., Global Positioning System [GPS], which operates in the 1164.1215, 1215.1240, and 15591.610-MHz frequency bands), has yet to be adequately validated (cf. Aiello et al., 2000; FCC, 2000). There is a second assumption that pulse signals above 2 GHz are relatively noninterfering due to propagation losses (FCC, 2000, p. 13, paragraph 27). Indeed, the effect of transient RF signals, as opposed to steady-state signals on materials and circuits, is a complex subject, but poorly understood (cf. Barrett, 1991, 1995a). Moreover, the effect of a train of aperiodic transient signals on conventional receivers and forms of electronic equipment may be a nonlinear temporal summation of the individual transients and a function of the relaxation time of a particular material or a particular circuit. Making the problem of interference even more complex is the fact that, although there is a short list of electronic materials, there is a long list of possible electronic circuits in victim receivers, each with a specific relaxation time regarding possible variables affecting susceptibility to interference. According to the FCC,

typical front-end bandwidths before the first mixer in receivers; typical dynamic range limits of receiver mixers; typical IF bandwidths; and required signal-to-interference ratios for reliable performance of the system assuming interference is white Gaussian noise . . .

FCC (2000, p. 14, paragraph 33)

These variables are, indeed, important. However, what is meant by “typical” is usually typical performance with respect to continuous, rather than transient, signals. Furthermore, a pulse transient is a broad spectral bandwidth signal (mathematically), but the frequencies are precisely phase-locked, not randomly phase-related as in white noise. Therefore, it is not clear that these “typical” measurements will provide an accurate prediction of interference by real transient signals. Despite this lack of knowledge concerning interference susceptibility, some UWB proponents do not believe in cumulative interference (cf. FCC, 2000, p. 21, paragraph 46).

In the case of more than one UWB communications system operating in asynchronous mode, the assumed absence of UWB-induced interference to other locally operating UWB systems is also yet to be validated. This form of interference, which may be absent or rare in the case of UWB radar, may yet be anticipated to be commonplace in the case of more widely used UWB communication systems.

2.7.2 Capacity of UWB Communications Systems

Shannon’s channel capacity laws are universally valid and apply to UWB communications systems, regardless of whether a government limits the bandwidth and the power used. However, UWB communications systems are yet to be evaluated with respect to both bandwidth efficiency and power efficiency. UWB communications systems’ bandwidth efficiency rating—the measure of bandwidth (i.e., real bandwidth, not merely the bandwidth that can be detected above conventional receiver thresholds) used together with data rate achieved in efficiency ratings—is presently extremely poor. UWB systems’ power efficiency rating—distance achieved for power (peak not average) used—is also poor. [A declared motivation of the FCC interest in considering permitting the operation of UWB systems is that it “would permit scarce spectrum resources to be used more efficiently” (FCC, 2000, p. 1)]. However, the aim to achieve efficiency addresses the issue of whether UWB transmissions do or do not interfere with the reception of conventional frequency receivers, that is, whether the noise floor of such receivers can be utilized without penalty. This is a different efficiency aim than the aim to achieve the highest data throughput through a channel of precisely defined and restricted bandwidth. Perhaps, it is not even an aim in efficiency. Engineering trades of time, bandwidth, and power assume a zero-sum game. Some proponents of UWB technology tacitly acknowledge the zero-sum game but claim that the S/N penalties from “reuse” of spectral areas already occupied by conventional narrow and broadband systems are spread over many victim receivers. Therefore, the argument goes, the penalty per victim receiver is small. Thus, rather than using the “scarce spectrum resources more efficiently,” operation of UWB communication systems would be an exercise in interference tolerance because there must be interference regardless of how little it is. But tolerance is not efficiency, and the amount to be tolerated is not trivial.

A UWB communications system is a strategy—a limiting case strategy—of utilizing a communications channel’s time-bandwidth-power product, bypassing the FCC bandwidth restrictions and allocating extremely broad instantaneous bandwidth to the symbol/signal. There are, of course, other strategies and other approaches to utilizing that same product but keeping within FCC guidelines. In the case of UWB radar/sensing, there are proven and demonstrated advantages for using UWB systems in precisely defined situations. However, the claimed advantages of the UWB communications approach are that—if noninterfering with other communications systems, which is more than doubtful—the approach provides modest data rates but robust communications in the presence of environmental interference factors and that the approach is superior in the presence of multipath transmissions. These claims may prove valid but are yet to be proven under normal operating conditions.

2.7.3 UWB Transmitters

A UWB communications transmitter system to a great extent shares the same systems configuration—if at lower power—as that of an electronic upset weapon or jammer. The differences lie mainly in the signal power levels at a set distance. As noted before, the transient response of a victim receiver or equipment is material- and circuit-dependent. The transition set of characteristics at which a noninterfering UWB communications transmitter becomes an electronic-upset UWB jammer is yet to be defined. Furthermore, once defined, it may be assumed that the transition set of characteristics will always be relative to the devices affected by the upset/interference. Transient effects are more complex than the steady-state effects. Therefore, regulatory rule-making will have to be complex.

In the recent past, and in the case of continuous wave systems, standards of emission have relied on PSD measurements. However, it is well known that the PSD measure, with its origins in harmonic analysis and with a relationship to the autocorrelation function, is an entirely inappropriate measure of transient and UWB signals. The PSD is an even function of frequency and possesses no phase information about the signal. A transient signal is not an even function of frequency, and a valid peak power measurement is critically dependent on signal phase. Some proponents of UWB systems have pointed to a low PSD as an indication of negligible interference potential with respect to narrowband receivers when, in fact, such a harmonic analysis is an inappropriate continuous wave (harmonic) analysis for a signal transient and the test instrument used to support such claims samples too slowly. A fast-rise-time pulse can produce multiple harmonic responses in a narrowband receiver, as well as considerable destructive heating effects.

The FCC publication of 47 U.S.C. Sections 154, 302a, 303, 304, 307, 336, and 544A (1), Subpart F—Ultra-Wideband Operation to Part 15, Radio Frequency Devices specifies the acceptable measurements of radiation from UWB devices (see Chapter 4).

2.7.4 UWB Radiation Measurements

The measurement of peak power levels is only as accurate as the sampling rate of the measuring device. It is worthwhile observing that a sampling rate is a measure of operations over time. Therefore, in assessing the peak power of a UWB transmitter, it is preferable to take the frequency bandwidth as being of secondary importance and focus on the signal duration and its rise time. If the reciprocal of the signal duration and rise time are greater than half the sampling rate of the measuring instrument (i.e., greater than the Nyquist rate), the measured power is not a true peak power measure. Yet some UWB proponents believe that the peak output is not the crucial variable in causing interference to a narrowband receiver but that only the PSD of the pulse and the PRF are causes of that interference (FCC, 2000, p. 19, paragraph 41). The FCC has proposed two methods of measuring peak power: the peak level of the emission over a bandwidth of 50 MHz and the absolute peak output of the emission over its entire bandwidth (FCC, 2000, pp. 19-20, paragraph 42). Of course, both proposals beg the question of how peak power is to be measured. The peak power in a 1-GHz monocycle signal measured by an instrument with a sampling rate of less than 2 GHz is actually an average power regardless of the emission bandwidth— instantaneous or sequential—sampled. Casting around for an appropriate measuring instrument, some faith has been placed in a pulse desensitization factor correction of an inadequately sampling spectrum analyzer (FCC, 2000, p. 23, paragraph 51, footnote 107), a method which guesses a true measure on the basis of measurement at an inadequate sampling rate (FCC, 2000, p. 24, paragraph 51). This method is clearly inadequate. All things considered, the (appropriately Nyquist-) sampling osci lloscope is probably an adequate measuring instrument (FCC, 2000, p. 53, paragraph 24) but only for the individual UWB pulse. The measurement of an asynchronous, aperiodic aggregate of UWB emitters can only be undertaken by real-time oscilloscopes such as the Tektronix 7000 series of DPOs, even if the aggregate could be assembled.

There have been claims by UWB proponents that a UWB signal will remain under the noise floor of a conventional receiver and thus provide no influence on that receiver’s performance. This claim is invalid. In the case of a single UWB emitter, the SNR in a victim receiver will be lowered because the noise floor will be raised. If many UWB transmitters are involved in aggregated networks, the noise floor of the victim receiver will be raised further. Thus, there is no escape from normal engineering trade-offs.

Time domain, not frequency domain, interference testing of UWB signals should be pursued, with (>20 GHz) sample-and-hold oscilloscopes and real-time DSOs for measuring individual pulse rise times and with DPOs for measuring aperiodic trains of aggregated emitters. Laboratory testing of UWB systems for interference is obviously only as accurate and as suitable as the laboratory equipment used. Laboratory testing with inaccurate and unsuitable test equipment provides invalid test results and supports invalid claims on performance. Simulation and analysis should support this laboratory testing.

Compounding this stew of unknowns is the UWB field in general and the development of UWB communications systems, which have been plagued by exaggerated performance claims in the public press, invalid priority and originality claims, and massive and ferocious legal and political activity. These circumstances amount to truly a unique phenomenon even by the standards of the times.

2.8.1 Overview

UWB radars/sensors also probe the target’s optical response more effectively, promise the capability of high-resolution target imaging and detection, and, in some application areas, have achieved commercial success in ground, wall, and foliage penetration; synthetic aperture radar (SAR); and target imaging. This section presents the pioneering, innovative, and fundamental contributions to the field.

2.8.2 Target UWB Response Aspects

A UWB radar or sensor system is perhaps more unconventional than a UWB communications system, not primarily in system components but in the physics involved, in most cases, the signal-target interactions. In the UWB radar case, the transmitted signal length cτ is shorter than the target. Conventionally, the complete radar cross section of any target can be specified by four complex numbers for a given frequency and set of orientation angles in a scattering matrix. In the case of UWB signals, this description of the radar cross section is still valid. However, the number of frequency components in the UWB transmitted signal is large, and different scattering components on the target will not only have different resonant frequencies but also, due to their spatial separation on the target, reflect the illuminating short-duration UWB signal at different times. Therefore, the UWB signal return is the sum of separate returns spread out discontinuously in time and consists of multiple time-separated scattering matrices, reflecting the separate spatial interactions on the target.

With respect to a conventional radar signal, which, in most cases is longer in spatial extent than the target, the target’s scattering matrix is not discontinuous in time. Therefore, with respect to most conventional radar signals, the target is a point scatterer—in contrast, in the case of UWB radar signals, the target response is decomposed into a collection of individual scattering components.

Using pulse compression techniques and chirped signals, conventional radars can also resolve a target into its individual scattering components, but only if the echo frequency difference Δf between, for example, two individual scattering components is equal to the reciprocal of the uncompressed pulse width τ or Δf = 1/τ.

Therefore, if the target subcomponent frequency differences are small, the chirp pulse leng th τ must be long. If τ is long, then the compression pulse width t_comp must be short. This forces the total change frequency over the duration of the uncompressed pulse Δf to be broad, according to the pulse compression ratio relation:

The shorter the compressed pulse, the broader the frequency change:

The requirement for a broad frequency sweep dictates a high change in transmitter frequency over the duration of the transmitted pulse τ. The practicality of high rates of change in the transmitter frequency is the delimiter for pulse compression techniques.

Thus, in the case of pulse compression, the frequency difference between the target subcomponent echoes must be large enough for the echoes to be resolved by a compression filter. UWB radars/sensors are able to resolve those subcomponents without the requirement of frequency difference because the returning echoes are already separated in time (nonoverlapping) before reception. Therefore, UWB radars/sensors offer an alternative method to high-resolution target imaging.

Moreover, depending on the ratio of the spatial/temporal length of the UWB signal to the length of the target, the UWB echo or target response can be one of at least three kinds: early-time (optical) response, resonance response, and late-time response (Cheville and Grischkowsky, 1995, 1997). Whereas the early-time response is target aspect-dependent, resonance and late-time responses are aspect-independent with respect to the harmonic components in the target’s response but are aspect-dependent with respect to amplitude of that response. Finally, each scattering component separated-in-time composing the complete UWB response of the target can possess, in varying degree, the three kinds of responses.

The first type of response component, the optical or early-time response, is only present in the case of a fast rise-time signal and only cleanly and commonly seen in the case of UWB radars/sensor signals. The resonance and late-time types of return signal components are also seen in conventional radar. The reson ance response of a target is established when the surface currents are present on the target or target components, and the late-time response occurs at the commencement of the decay of the resonance response and after the end of the exciting signal.

The optical or early-time response is more associated with UWB radar. The radar community was first made aware of this target response by Morgan (1984) and Pearson (1984), who pointed out that a forced component—in addition to the damped sinusoidal components (the resonance and the late-time response)—is an essential part of the scattered response over the time interval during which an impulsive plane wave is present on the scatterer (see also Auton and van Blaricum, 1981; Baum, 1971, 1976; Kennaugh and Moffat, 1965; Moffat and Mains, 1975; van Blaricum, 1978, 1991; van Blaricum and Mittra, 1975). This means that a transient scattered field cannot be expressed purely as a harmonic response until the scatterer’s natural modes are established, that is, until the resonance component is established. The early-time (optical) response of the transient scattered field is characterized by time-varying coefficients determined by local features of the scattering object and is due to direct physical optic fields, as well as a sum of temporally modulated natural modes. The duration of this early-time response is also equal to the time the wave shape is present on the scatterer. In some instances and orientations, the early-time (optical) response components of the return signal can be of much larger peak amplitude than the resonance and late-time responses.

Radar signals can be characterized not only with respect to duration and spatial extent with respect to the target but also with respect to the target size length/signal wavelength ratio (L/λ). Wavelength and target-size dependent scattering regimes are as follows: the Rayleigh region, where the signal wavelength is much longer than the size of the target (L/λ ≪ 1) and the reflected signal is inversely proportional to the fourth power of the wavelength; the Mie or resonance region, in which the wavelength of the signal is of the same order of magnitude as the size of the target (0.5 < L/λ < 10); and the optical scattering regions, in which the wavelength of the signal is much shorter than the target (L/λ > 10), and, in the case of some target shapes, the target reflectivity can be frequency independent. UWB radar return studies address the scattering regimes of the resonance and optical regions, and the three return signal components discussed above fall in those regimes.

2.8.3 Radars and Sensors

There are marked differences between a conventional and a so-called UWB radar/sensor, that is, an ultrashort-pulse radar/sensor, with respect to signal design, emitters, receivers, and forms of processing. There are also design differences between a UWB radar and a UWB sensor. In fact, the UWB radar designer must decide up-front whether to design a UWB radar or sensor, both of which are different from conventional radars. The purpose of the UWB radar is to provide target detection, whereas the UWB sensor provides information concerning the target, that is, target identification. Thus, the UWB radar has more relaxed receiver requirements than a UWB sensor.

Despite the unfortunate UWB labeling, the fundamental distinguishing feature of the UWB emitted signal is not the extreme bandwidth, but that it is ultrashort in time and spatially shorter than the length of most targets. The wide bandwidth of a UWB signal derives from its ultrashort duration. The fact that the UWB signal length is shorter than the length of a target means, as previously explained, that the target’s response decomposes into its individual scattering components and that those returning signal components are spread out discontinuously in time. In the case of conventional radars, the signal is usually longer spatially than the length of the target, which means that the target response is not spread out discontinuously in time. A target irradiated by a UWB signal becomes a collection or aggregate of point scatterers, all of which are associated with subcomponents of the whole target. Thus the term UWB radar pulse (or ultrashort RF pulse) can only be adequately defined relative to a stated target spatial extent L. Considering a pulse that is ultrashort, and therefore UWB, with respect to a Boeing 747 may not be so with respect to a cruise missile or even with respect to a subcomponent such as an engine on a Boeing 747. Presently, there is no agreed absolute duration, bandwidth, or Q defining a UWB signal. In this article, and in the definition of UWB technology, the applicable advice (Taylor, 1995a, p. 3) is “because ultra-wideband is a new term, it is best to look for the writer’s definition or to determine the meaning in context and the accompanying details, and then apply the mathematical descriptions loosely.”

Some helpful definitions include the following:

However, not all definitions of UWB radar based on the above definitions are useful. An OSD/DARPA (1990) panel provided a UWB signal definition using fractional bandwidth:

Ultra-wideband radar is any radar whose fractional bandwidth is greater than 0.25, regardless of the center frequency or the signal time-bandwidth product.

However, in discarding any reference to the duration of the pulse from the definition, the OSD/DARPA panel essentially overgeneralized the label, permitting it to apply to any signal that occupied a fractional bandwidth greater than 0.25, regardless of the time over which that occupancy occurs. Thus, a chirped signal, the instantaneous bandwidth of which is narrow, but which occupies over 1 s or more a fractional bandwidth greater than 0.25, would be considered UWB, according to the panel’s definition. Selection of a 0.25 fractional bandwidth boundary is, of course, arbitrary; the panel’s definition captures neither the ultrashort nature of the “UWB” signal nor the ultrawide instantaneous bandwidth resulting from that extreme shortness. Nevertheless, the label “UWB” has been attached to this technology, and it is perhaps not too late to remove it. (The laser physics community was not so plagued by inappropriate labels. Ultrashort radiation, and hence UWB radio and light pulses, are called ultrashort pulses—cf. Hopkins and Sibbert (2000).

If one conceives of the target reflection of an emitted signal and the return of the transformed signal to the emitter in terms of a signal input to a system and signal output from a system, then the reflectance properties of the target are described by a transfer function (in the frequency domain) or impulse response (in the time domain). In these system theory terms, a target’s transfer function in the case of a conventional radar is usually, at steady state, a unitary response function of a duration associated with target size; however, in the case of UWB radar, it is a time series of aggregate responses associated with the size, orientation, and material composition of the target subcomponents. In fact, the UWB emitted signal, target reflectance (transfer), and signal reception can be considered an informational channel, in which intelligence about the target is impressed on the emitted signal by the target at the commencement of that channel, and that intelligence or information arrives at the receiving antenna or at the end of the channel.

Given this intelligence about the target return arriving at the receiving antenna, the design engineer has the choice of either trading the received intelligence into a high-signal-to-noise (S/N) detecting radar or preserving that intelligence (baseband signal) in a sensor capable of target identification, characterization, and imaging.

2.8.4 UWB Receivers

Because of the UWB signal’s fast rise time, the UWB receiver must have appropriate bandwidth. The receiver must have a sampling rate adequate to preserve the peak power against averaging or integration. Detection of the signal then proceeds by either threshold or correlation methods.

Taylor and Kisenwether (1995) contrast the narrowband receiver design bandwidth requirement based on the pulse duration τ as

for which, in the case of conventional pulse detection, the detector section must operate up to the frequency

while an UWB detector must operate up to an upper frequency

where t_r is the pulse rise time. In fact, the UWB receiver detection circuit is driven by the pulse’s rise time and not the pulse width. Note that the time derivative of the receiving-antenna forcing function ∂i/∂t, where i is the receiver current in amperes and E is the electric-field strength (volts/meter) of the received signal before reception, is such that ∂i/∂t = E.

The increased bandwidth associated with the received signal rise time increases the noise in the receiver, and because UWB signals have a wide relative bandwidth, signal components at the low and high bounds are processed with different fidelity, dispersion, and loss.

Taylor and Kisenwether (1995) point out that, whereas in designing conventional receivers, it is usual to describe the signal in units of volts (J/I × s), or watts (J/s), in the case of UWB receivers the signal is conceived in terms of volts × second (J/I), or watts × s² (Js). These signal definitions reveal that there is a clear requirement for the UWB receiver to measure the signal energy and not the average signal energy. In other words, there is a requirement to measure the signal peak energy or power, not the average signal energy or power.

As stated previously, the designer of UWB radar systems can choose between designing a receiver for a target-detecting UWB radar or a target-discriminating and imaging UWB sensor. If the intention is to design an imaging UWB sensor, then the waveform of the UWB return signal must be preserved; however, in the case of a target-detecting UWB radar, it need not be preserved. In the case of a UWB sensor, a homodyne receiver is appropriate, and in the case of a UWB radar, either a homodyne or a superheterodyne receiver is appropriate (cf. Barrett, 1995b; Taylor and Kisenwether, 1995).

Detection can be performed by threshold detection or correlation detection (with a requirement of a SNR at least 10 dB). In the detection stage of a UWB radar, nonlinear threshold detection is adequate (cf. Robbins, 1972), but in the detection stage of a UWB sensor, the requirement to preserve the original return signal waveform requires sufficient bandwidth associated with expected signal rise/fall times, that is, preservation of the modulations impressed on the signal by the target.

Taylor and Kisenwether (1995) showed that for threshold detection in the case of the conventional narrowband receiver, the performance degrades (minimum detectable signal [units: watts or volts] is greater) as the receiver bandwidth increases because of increased broadband noise.

In the case of the UWB receiver, the performance improves (minimum detectable signal [units: watts × s² or volts × s]) as the receiver bandwidth increases because more signal transient energy is captured and because of the assumption that the signal bandwidth is greater than the noise bandwidth (if the signal bandwidth is not greater than the noise bandwidth, then this argument would need amendment).

Therefore, whereas the design objective for conventional, narrowband receivers is to decrease the bandwidth to improve narrowband signal sensitivity by rejecting impulse signals, the design objective for the UWB receiver is to maximize the bandwidth to improve impulse signal sensitivity, and thereby neglect frequency selectivity.

If there is no requirement to preserve the waveform of the UWB return signal, that is, the case of UWB radar, not sensor, then correlation techniques can be used for detection. There are two forms of correlation detection of the individual pulse, autocorrelation and cross-correlation:

There is extensive literature on the processing of pulse trains. It is well known that the PSD of a pulse train is proportional to the absolute value of the Fourier transform of the single pulse. This despite claims by some UWB proponents that changing the interpulse intervals in a pulse train, or dithering, flattens all of the power density spectrum. In addition, the phase information in the Fourier transform of the individual pulse is not retained in the autocorrelation; therefore, the power density spectrum of the pulse train does not have a unique solution (i.e., the same PSD can be obtained from different collections of signals).

Neglecting these drawbacks and having obtained the PSD of a pulse train, it can be used in various ways. For example, if the pulse train is impressed with Barker or complementary (Golay) codes, then the instantaneous power of each pulse can be reduced to achieve the same total energy interaction with the target over the pulse train. This assumes that the target is not performing maneuvers over a timescale comparable to that of the pulse train and that the target’s response is not extremely aspect-dependent. Also, time and spatial resolution can be improved by pulse compression over the length of the pulse train using well-known methods. In addition, the SNR of the return signal can be improved by the same pulse compression methods.

It should be noted that, given the intention to design a fast UWB receiver preserving the modulations impressed by the target on the signal, the heavy continuous processing burden requiring 20-GHz sampling rates must be addressed. For example, these sampling rates force the designer to consider receiver designs based on digital phosphor and SiGe technology.

2.8.5 Ground-, Wall-, and Foliage-Penetrating Radars

UWB radars for ground and wall penetration offer the obvious advantage of not requiring simultaneous transmission while receiving from close proximity targets. Almost all ground-penetrating radar (GPR) systems presently operate at or below 1 GHz primarily because the attenuation of the signal by the ground increases with frequency (see Daniels et al., 1988; Elachi et al., 1984; Pittman et al., 1982). However, two other mechanisms may also be operating, that is, the ground can also be a dispersive medium, and therefore, the lower frequencies in the return signal at the surface can either be due to low-pass filtering or medium dispersion, and the relaxation times of various ground media are comparatively long (Barrett, 1991, 1995b), which means short-pulse envelope effects can occur. The relative influence of these different mechanisms is unknown at the present time.

The presence of water in the ground, which can influence ground conductivity, is important because, according to Maxwell’s theory, dielectric constant and conductivity always occur in combination, and thus together they will affect the propagation of waves in matter (cf. King and Smith, 1981). The phase shift in a propagating wave is related to both the medium’s effective permittivity and effective conductivity (Ulriksen, 1981a and 1981b, 1982). Furthermore, the relaxation time of water in soils is displaced to a lower frequency than that in bulk water (Hoekstra and Delaney, 1979), and there appears to be two closely spaced relaxation times, rather than one (De Loor, 1983).

In GPRs, a number of techniques are followed to remove noise and artifacts (Osumi and Ueno, 1988; Safar, 1985; Schneider, 1978; Yeung and Evans, 1988), a major artifact being due to the footprint size of the beam projected from a moving vehicle being cone shaped and wide so that there are multiple interactions with a single stationary below-the-surface target, hence multiple overlapping spatial returns.

The UWB signal, which delivers a wide bandwidth in a short time, is compatible with the requirements for a SAR (Lahaie, 1992). The UWB signals from aircraft equipped with UWB SARs can penetrate foliage (FOPEN) to reveal objects concealed in forest and vegetation (Vickers, 1992) by utilizing the capability of UWB pulses to deliver a wide bandwidth in a short period of time.

2.8.6 Russian UWB Radar Systems

There has been extensive development of UWB systems and subsystems in the former Soviet Union and the present Russian Federation. This work arose out of programs to improve power systems in the 1950s. Even at this stage, the difference between conventional continuous wave signal description methods and ultrashort-pulse methods (e.g., Zernov, 1951) was noted. The simplicity of the methods of time domain analysis for short-pulse UWB signals, as opposed to continuous, steady-state signals, was described by Kharkevitch (1952).

Initially, radio pulses of nanosecond duration were generated using traveling wave tube modulation (Astanin and Kardo-Sysoev, 2000). In 1957, Astanin, while at the A. Mozjaisky Military Air Force Academy, developed an X-band 0.5-ns duration transmitter for waveguide study. A receiver-correlator with a T-bridge waveguide and mechanically controlled delay was used (Astanin, 1964). At the same time, at the Radioelectronics Institute of the USSR Academy of Science, Kobzarev and collaborators conducted tests on indoor ranges of ultrashort-pulse high-resolution radars. These constituted the first stage of development of UWB systems in Russia (Astanin and Kardo-Sysoev, 2000).

The next stage of development utilized fast semiconductor switches, beginning with Shatz (1963) and continuing at the Ioffe Physico-Technical Institute. As in the United States, progress was facilitated by the availability of fast-sampling oscilloscopes—initially at Novgorod (Rjabinin) and at Vilnius, Lithuania (Efimchik and Levitas, 1976). The theoretical basis for timescale transformation procedures was developed by Naidenov (1978), and eventually 10-GHz sampling oscilloscopes were developed. At this time, the applications were in GPR (Finkelstein, 1994; Shirman, 1991). It was realized that UWB signals constitute a separate class of signals and that Doppler, in the case of ultrashort-pulse trains, becomes not a measure of phase but a timescale transformation (Astanin and Dorsky, 1988; Astanin and Kostelev, 1989). These investigations resulted in the formulation of a time domain analysis of signals, which is discussed in Radar Target Characteristics: Measurements and Applications (1994) describing studies involving target impulse response, ramp response, as well as signal shape and signal structure description (Astanin et al., 1994).

While the different nature of ultrashort pulse or UWB systems from conventional radars became enveloped in rancor and controversy in the United States, it was first realized in Russia that ultrashort (shorter than target length) pulses deliver more target parameter information, such as target state and orientation, than do conventional long (longer than target length) pulse systems. This realization led to the addressing of target classification and imaging, as well as to the theoretical issues of ill-posed problem solutions (Kostylev, 1984; Tikhonov and Arsenin, 1986). The theory of ill-posed problems became a new branch of mathematics, and its basic results were first obtained by Soviet mathematicians. Treatments of ultrashort-pulse signal returns also used tomographic methods (Kononov, 1992) and wavelets (Astanin and Kostylev, 1997). The development of ultrashort-pulse transmitters even included picosecond and femtosecond pulse transmitters (Glebovich et al., 1984). In recent years, there have been major theoretical advances (e.g., Astanin et al., 1994; Borisov and Utkin, 1995; Feld, 1991), as well as advances in mine detection (Astanin et al., 2000).

From a subcomponent standpoint, the obvious solutions were used, as they were in the United States. For example, Astanin and Kostylev (1989) summarized the earlier work addressing a time domain transmitter and receiver and observed, “Generators based on avalanche transistors are widely used.” Meleshko (1987), in a section titled “Time Correlation of Pulsed Signals” states, “In devices which implement the first, simplest method, correlation is performed at the moment when the leading edge of the input pulse crosses the constant threshold.” Furthermore, many circuit designs of tracking-threshold shapers are shown, indicating that the use of signal edge detection was commonplace.

Varganov et al. (1985, p. 5) provided one definition of superwideband signals, namely Δf/f = 1 and described a number of methods for transmitting and receiving impulse signals. Presently, there are a variety of definitions of UWB including IEEE STD 1672. According to OSD/DARPA (1990), UWB devices must have a 20-dB fractional bandwidth of at least 0.25, where the fractional bandwidth is 2(f_H – f_L)/(f_H + f_L), where f_H is the upper frequency of the 20-dB emission point and f_L is the lower frequency of the 20-dB emission point. The center frequency is defined as

The FCC adopted the OSD/DARPA (1990) definition, but with a modification: UWB devices are defined as any devices where the fractional -10 dB bandwidth is greater than 0.20 or “occupies 0.5 GHz or more of spectrum.” The FCC also provisionally proposes to base the UWB definition on the 10-dB bandwidth, rather than the 20-dB bandwidth. Furthermore, the FCC adopted the OSD/DARPA (1990) definition of center frequency (FCC Subsection 45, Ultra-wideband Radiation).

In the late 1960s and early 1970s, Kardo-Sysoev discovered a new method called avalanche injection (AI) for switching modular thyristors. Even at these early dates, switching times of ~10 ns at 1 kV and >100 A were achieved (Kardo-Sysoev et al., 1976). The group at the Ioffe Physico-Technical Institute, St. Petersburg, headed by Kardo-Sysoev, also pioneered the development of delayed ionization switches called silicon avalanche shapers (SAS) (Grekhov et al., 1981). The SASs required high rise time triggering (>10¹² V/s) and have been used as shaping heads for thyratrons and vacuum tubes. A second generation of devices called drift step recovery diodes (DSRD) were capable of supplying that triggering (Grekhov et al., 1985). These developments made possible solid-state pulsers with greater than megawatt peak powers and 0.1-ns fronts. The devices were used in the late 1980s in the development of an antistealth UWB pulsed phased antenna array, which was ultimately cancelled on the collapse of the USSR (Kardo-Sysoev, personal communication, 2000).

Presently, the Ioffe Physico-Technical Institute group offers an extremely powerful pulser combining all the earlier works in the form of compressor cells. The compressor cells are modulator thyristors for 100-ns cells, DSRDs for nanosecond cells, and SASs for subnanosecond cells. Russian GPRs include those of I.M. Finkelshtein (Riga) and V.E. Kotenkov (Moscow).

2.9.1 Singularity Expansion Method

The singularity expansion method (SEM) estimates the target scattering matrix in Laplace transform form. As is well known, the Laplace transform of the target’s impulse response, that is, its transfer function, can, in many cases, be represented or summarized in terms of the poles and sinks of the function. This pole and sink summary can provide a signature permitting target identification. The presupposition of the approach is that the target’s transfer function is linear—which it generally is, but not always. (Note that the white noise analysis approach does not assume that the target’s transfer function is linear. In fact, this approach is a method for detecting nonlinearities.) The great advantage of Laplace analysis is that the transient, as well as the harmonic, nature of the target’s transfer function can be characterized. Van Blaricum (1995) discussed the history of the development of the SEM approach.

The SEM method can characterize the target response in terms of the optical, natural resonance, and standing waves. The resonance poles of a target are aspect-independent, but the ability to couple to them is not. If the SEM application is applied predominately to the natural resonances across aspect angles, the result can still be encouraging, but the method requires close attention to using a processing method that addresses the return signal components from target resonances excitable at all angles and neglects those resonances excitable only at certain angles.

2.9.2 White Noise Analysis

The Wiener functional expansion is a general method for analyzing nonlinear systems that have a finite memory. A system is nonlinear if the principle of superposition does not apply to the system, either because the system’s frequency aggregate dispersion response is not a linear additive function of the individual single-frequency dispersion responses (dispersive nonlinearity) or because the system’s aggregate amplitude response is not a linear additive function of the individual single-pulse amplitude responses (intensity-dependent nonlinearity).

In the case of pulsed excitations, a system’s nonlinearity can be described as an influence of the past input to a system on the response of the system to a present input to the system, which attenuates with time, that is, the system has memory. For example, the late-time response of a target exhibits a form of memory of the driving signal no longer present on the target. The decay time or offset time of a receiver is also an indication of this form of memory. Thus, there is an interpulse temporal threshold, that is, the driving pulses must be separated less than a certain interpulse interval Δt for a memory influence to occur. However, after that threshold is crossed (Δt < threshold), the response of the system adds. The addition can be instantaneously linear, that is, the principle of superposition can apply to the instantaneous addition, but due to the aforementioned temporal threshold and the response decay over time, the system is nonlinear according to the previous definition, that is, the principle of superposition does not apply over time. There is no system response addition when the interpulse interval Δt is greater than the threshold and the amount of addition of the first response to the second response is a function of the system’s decay time. This kind of system response is sometimes called temporal summation.

With those definitions in mind, methods due to Volterra and Wiener, which underpin white noise analysis, are explored here. Volterra (1959) showed that the transfer function or impulse response of a nonlinear, time invariant, finite memory, analytic system can generally be characterized by a generalization of the Taylor series of multiple arguments known as the Volterra series. Next, Wiener (1942, 1958) and Lee (1964) showed that if the input to an unknown system is Gaussian white noise and if the Volterra series components are mutually orthogonal, it is possible to characterize the Volterra description of a system in terms of a series of kernel functions, now known as the Wiener kernels. Wiener constructed a hierarchy of these functionals (or functions of functions) that are orthogonal to each other with white noise as the stimulus to the system under consideration. The Wiener series representation of a system’s transfer function/impulse response is related to the Volterra series; however, as it is an orthogonal series, it has a stronger convergence than the Volterra series. Once the set of Wiener kernels are estimated, a system’s transfer function/impulse response is completely characterized. The Wiener kernels were difficult to compute until Lee and Schetzen (1965) showed how these kernels can be evaluated by well-understood cross-correlation techniques.

White noise analysis methods are general and describe the transfer function/impulse response of any unknown system (cf. Barrett, 1978; Marmarelis and Marmarelis, 1978). More recently, these methods have also been applied to characterize the transfer function/impulse response of targets to RF signals.

An approach derived from white noise analysis is bispectral analysis (Jouny and Walton, 1991; Marmarelis and Sheby, 1988), which involves triple autocorrelations and can detect return signals with skewed amplitude probability distributions. Another derived approach is that of (Wiener) kernel analysis (Marmarelis et al., 1995) involving the characterization of the target transfer function/impulse response in terms of higher order functionals (convolutional integrals) or Wiener kernels. The higher order kernels, higher than the first, describe the system nonlinearities, and the full set of kernels constitutes a generalized impulse response function. The transmitted signals are based on pseudorandom or continuous-time broadband random processes with zero mean (Marmarelis et al., 1995). The technique permits noise suppression, provides phase information, and also signal reconstruction. Preliminary results characterizing materials and targets using these techniques have been promising.

2.9.3 Time-Frequency Approaches to High-Resolution Target Identification

A strict adherence to merely time or frequency methods alone removes from consideration powerful methods for characterizing and identifying targets. These new methods fall in the general category of time-frequency methods. If a signal is decomposed into a time-frequency wavelet or wavelet packet representation, and multivariable correlations (e.g., across time or frequency) are performed on that two-dimensional representation, then signal response differences between targets are made explicit and can be quantified by correlation.

To illustrate, two representative UWB target return signals are shown in Figure 2.12 and in a time-frequency representational form in Figure 2.13, and frequency-frequency correlations of the time-frequency matrices are shown in Figure 2.14. (Values across the diagonal are identical. Values along the diagonal are unity.) Figure 2.15 shows correlations of the time content of the frequency bins and covariances. The correlations of set frequency components of the two return signals over time are low and provide a characterization of differences between the two targets’ instantaneous frequency response over time. These data provide indication of time-and-frequency differences in the signal return from the two targets. Each target return signal is shown to have a distinctive signature or fingerprint not apparent in the raw signal. In the case of these particular UWB return signals, the frequencies at set times are correlated negatively at 0.025, but the times of set frequencies are correlated positively at 0.67. After this processing, the difference between the two signal returns can be easily distinguished, as shown in Figure 2.15, and the two UWB target signal returns, which at first appeared very similar, show a small correlation in most instances over the analyzed time-frequency plane. Thus, the time-frequency decomposition processing provides a powerful method for making explicit discriminating differences between target signal returns and permits target recognition and identification on the basis of pattern. Figure 2.16 provides the frequency-frequency cross-correlations of the individual time-frequency plots for the Cessna and F-4.

2.9.4 Time-Frequency Approaches to High-Resolution Target Detection

The high-resolution target detection capabilities of UWB radars/sensors have advantages and a related disadvantage. The disadvantage is that a transmitted UWB pulse will interact with a variety of targets. Therefore, signal returns from targets of no interest will crowd the receiver as undesired interfering noise, or clutter, lowering the designated target’s SNR. Of course, what is designated clutter and what is designated a target are matters of choice. Furthermore, the choice can change. Nonetheless, given a designated target, clutter lowers the SNR of the return signal from the chosen or designated target.

A method for enhancing the designated target SNR in the presence of designated clutter, which also permits changes in the choice of designated target and clutter, was suggested by Barrett (1996)—the matched pulse or matched adaptive time-frequency packet (MAP) signal approach. The MAP method is a time-frequency approach and requires a priori information about the impulse response of the designated target or classes of target.

Figure 2.17 shows the general MAP procedure. Suppose a requirement exists to enhance the SNR selectively for two targets or classes of targets: target 1 and target 2. Further, suppose that the targets 1 and 2 will also serve to provide the target/clutter signal return for each other. That is, the choice of which is designated target and which is designated the clutter can be changed. (Figure 2.17a and b) using a UWB signal, for example, a monocycle, as the transmit pulse. The received signal in each case is treated as the impulse response of each target. These received signals are then time reversed to form their complex conjugate, and for purposes of exposition, they are called matched pulse 1 and matched pulse 2. Information concerning a target’s impulse response and thereby of its matched pulse constitutes a priori information about a target or class of targets.

In a second operational scenario (Figure 2.17c and d), the matched pulse for target 1 is transmitted and the return signal from targets 1 and 2 is received. Neglecting issues of the scattering matrix’s polarization and aspect dependence, according to systems theory the return signal from target 1 is enhanced (Figure 2.17c), and depending on the relative orthogonality of impulse response of target 2 with respect to that of target 1, the return signal from target 2 is diminished (Figure 2.17d). That is, the matched pulse 1 enhances the SNR of the signal return from target 1.

Conversely, in Figure 2.17e and f, the matched pulse for target 2 is transmitted and the return signal from targets 1 and 2 is received. Again, neglecting issues of the scattering matrix’s polarization and aspect dependence, according to systems theory the return signal from target 2 is enhanced (Figure 2.17e), and depending on the relative orthogonality of impulse response of target 1 with respect to that of target 2, the return signal from target 1 is diminished (Figure 2.17f). That is, the matched pulse 2 enhances the SNR of the signal return from target 2.

This method can be utilized to reduce clutter with resonances different from those of the target, to discriminate otherwise very similar targets or classes of targets on the basis of small differences between the targets’ scattering matrices, and to provide early warning of known threatening targets or classes of targets.