Antennas

An antenna is a device designed to accept RF power from a transmitter and radiate it into its surroundings, or alternatively to extract energy from a passing radio wave and deliver it to a receiver. Considering transmitting first, an antenna is ideally designed to present a resistive load R_t = R_r + R₁ (a pure resistance equal to the design load impedance of the transmitter, usually 50 Ω, if perfectly tuned and matched) and it is to this resistance that the transmitter delivers power. If the antenna is also loss-free, all the power delivered to it goes into the radiation resistance R_r and is radiated; if not, a proportion of it is converted into heat in the antenna’s loss resistance R1. The efficiency η of an antenna is given by η = . An ideal isotropic antenna is loss-free and radiates power in all directions with an equal intensity; it is a figment of the imagination as Maxwell’s equations describing electromagnetic radiation do not permit of such a design, but it is a useful yardstick for practical antennas.

Practical antennas fall into two main groups, those which are self-resonant and those which are not. But note that in use, non-resonant antennas are often brought to resonance, e.g. with the aid of an ATU ‘antenna tuning unit’. The simplest resonant antenna is the half-wave dipole (known in the Americas as a doublet), the fields in the vicinity of which are shown in Figure 13.1. Figure 14.1a shows its figure-of-eight vertical radiation pattern in cross-section. The radiation intensity is a maximum in the plane at right angles to the dipole and is ‘doughnut’ shaped; there is no radiation along the line of the dipole. A vertical dipole is described as ‘vertically polarized’ since the lines of electric field in the direction of maximum radiation are vertical. As can be seen, the two halves of the figure eight are not quite circular. They are exactly circular for a dipole very much shorter than half a wavelength, but such an antenna is not resonant. In the direction of maximum radiation, the field strength produced by a lossless resonant dipole is 1.28 times that of an isotropic radiator, or ‘2.15 dB above isotropic’, whilst for a (suitably-matched loss-free) short dipole it is 1.22 times (1.76dB). When considering a perfectly-matched lossless dipole, these figures also represent the ‘directivity’ or gain relative to an ideal isotropic antenna. However, the term ‘gain’ should be restricted to the ratio of the actual maximum field produced by an antenna, relative to that which would be produced by an ideal isotropic antenna, i.e. ‘gain’ takes into account an antenna’s losses due to R1. Only in the case of a perfectly-matched lossless antenna does the directivity equal the gain in the maximum direction. In the case particularly of antennas which are not self-resonant, the difference between gain and directivity can sometimes be very large, even when the antenna is brought to resonance by tuning.

Due to end effects, a thin wire radiator such as that in Figure 14.1a has an electrical length which is about 0.025λ longer than its physical length. Like all resonant circuits, a resonant antenna has a bandwidth depending upon the circuit constants. For thin wire dipoles - length/diameter of the order 500:1 - the useful bandwidth for transmitting is about +/-10%, limited by the increase in VSWR away from the resonant frequency; rather more for receiving, where a worse VSWR is usually acceptable.

The bandwidth of a dipole can be increased by making the conductors very fat - tubes or wire cages - over most of their length, tapering conically to the feedpoint. A variant on this theme, the discone antenna, is illustrated in Figure 14.2. The operating frequency range may be increased if an ATU (antenna tuning unit) is used to bring the dipole back to resonance. The ATU actually decreases the ‘instantaneous bandwidth’, but the ATU can retune the dipole to resonance when a different operating frequency is required. For very broadband signals, the instantaneous bandwidth of an antenna can be increased by a technique known as compensation [1]. The impedance of a centre-fed λ/2 dipole (Figure 14.3a) is low and resistive, typically 73 Ω balanced. To generalize, it is low for dipoles an odd number of half-wavelengths long, and high for an even number of half-wavelengths (e.g. Figure 14.1c and b respectively) as is clear from the current distributions. For other lengths the impedance is not resistive; such dipoles are not resonant. The radiation patterns for dipoles having lengths of multiples of the half-wavelength at the operating frequency show additional lobes, e.g. for lengths 1 and 1.5 times the wavelength (see Figure 14.1). Note that the number of lobes is equal to twice the number of half-wavelengths. The patterns shown are for antennas in free space, i.e. remote from the ground, which would act as a reflector and modify the patterns.

The 73 Ω impedance of the half-wave antenna of (Figure 14.3a is not convenient for connecting to a balanced twin wire feeder, which usually has an impedance of about 300 Ω, but this can be accommodated with a ‘delta match’ (Figure 14.3b). On the other hand, 75 Ω coaxial cable is about the right impedance for direct connection, but is unbalanced. A 1:1 ratio balun transformer (see Chapter 4) could be used, but this is a broadband device which is rather a waste as the dipole is inherently a narrow band radiator. A narrow band balun can be realized in various ways as in (Figure 14.4, and with proper choice of dimensions can also match the antenna to a 50 Ω cable, this impedance being preferred for transmitting systems. For receiving, e.g. for UHF Band IV/V TV, 75 Ω coax is commonly used without a balun, the balanced to unbalanced transition taking place gradually over a distance of several wavelengths along the feeder. Note that a wavelength in the cable is only about 0.6λ, as the velocity of the signal in the cable is only about 60% of that in free space. For VHF FM, a balanced 300 Ω twin wire feeder is often used and here the folded dipole of Figure 14.3c is useful. The two close-spaced dipoles act as a 2:1 turns ratio transformer, transforming the 73 Ω impedance of the simple λ/2 dipole to 292 O. A feeder which passes close to a source of interference is less prone to pick-up if it is balanced; in the case of an unbalanced feeder, an interference voltage may be induced in series with the outer, dividing (not necessarily equally) between the antenna and the receiver. In the case of a balanced feeder, the interfering voltage is induced equally in both conductors of the pair as a common-mode or ‘push-push’ signal, whereas the receiver (ideally) only responds to the normal mode (transverse or push-pull) voltage between the conductors. Incidentally, a folded dipole is often used in a Yagi multi-element antenna, connected to a 75 Ω feeder. The explanation is that one effect of the parasitic elements (reflector and directors) is to greatly reduce the impedance of a simple λ/2 dipole: using a folded dipole restores the desired 75 Ω impedance level.

The antennas which have been considered so far are balanced types. The operation of unbalanced antennas can be approached by looking at the performance of a modified balanced antenna. (Figure 14.5a shows a vertical λ/2 dipole with a horizontal metal sheet of very high conductivity and infinite in extent (a copper sheet extending many wavelengths would be an adequate approximation) inserted between the two halves, and its equivalent circuit. Note that the electric lines of force all meet the metal sheet at right angles and so are unaffected, whilst the circular horizontal magnetic lines, being parallel to it, do not cut the conductor and so are also unaffected. Therefore the field pattern is likewise unaffected, half the power being radiated above the plane and half below. If now the lower dipole element is removed and all the power fed into the top element (taking care to match the altered input impedance of 37 O), the far field of a λ/4 monopole above a conducting plane is seen to be the same shape as the upper half of the pattern for a λ/2 dipole but 3 dB higher in strength, or 5.15 dB above isotropic. The conducting plane is usually a ‘ground plane’, e.g. soil of very good conductivity. If the ground plane is not perfect (e.g. normal soil conditions) then the main lobe does not extend down to ground level. This is shown dotted in Figure 14.5b for the case of a λ/4 monopole (but applies equally to the other patterns), and the VSWR of the antenna will be high. The VSWR can be greatly improved with a set of buried radial conductors or a chicken-wire earth mat extending out to a radius equal to the antenna height, but for any significant improvement in the low angle radiation the mat would need to extend so much further that it is usually not economic so to do. Figure 13.5b shows the case of various monopoles including top loaded λ/4 monopoles (T and inverted L, useful to minimize antenna height when the wavelength is long), and the monopole. Monopoles up to λ/2 high have only the main lobe, which comes down to ground level; at λ small secondary lobes appear and at these are as large as the lower lobes. (Note that the descriptions T and inverted L are usually applied to antennas which are very much shorter than λ/4 and consequently not self-resonant even with the top loading, and must be brought into resonance by inductive loading. Medium and long wave broadcast antennas are of this type. Here, the top capacity loading is used to bring the effective height of the antenna closer to the physical height.)

In the case of an antenna elevated above ground, the situation is more complicated, the radiation pattern in the vertical plane depending upon the pattern of the antenna itself, its height above the ground plane, its polarization, and the nature of the ground. Horizontally polarized waves suffer a phase reversal on reflection, exactly so and without loss if over a perfect ground plane. Thus there may be considered to be an ‘image’ antenna below ground, energized in antiphase. Since all points at ground level are equidistant from the antenna and its image, there is no net radiation at zero elevation. Vertically polarized waves are not phase reversed at angles above the ‘peudo-Brewster angle’, but are phase reversed below it. For perfect ground, this angle is zero, giving a maximum of radiation at zero elevation angle. But in practice, with normal or even ‘good’ ground, the peudo-Brewster angle is not zero, so that for rays at grazing incidence, there is phase reversal on reflection and hence a null at zero elevation.

In the case of either horizontally- or vertically-polarized antennas, the radiation pattern in elevation may exhibit one or more lobes, depending upon the antenna height above ground. The greater the height (in wavelengths) of the antenna above ground, the more lobes will appear. On the other hand, the horizontal plane or azimuth pattern depends upon that of the antenna itself, so for the vertical dipole it will be omnidirectional, and for the horizontal dipole basically figure-of-eight.

Many investigations of the radiation patterns of various antennas have been carried out, both in simulation and by actual measurements (e.g. by overflying by helicopter fitted with a measuring antenna). Given the many different types of antenna, varying mounting heights and allowing for the wide range of frequencies used for communications, the possible permutations are infinite. (Figure 14.6 shows a computer-simulated radiation pattern of a horizontal half-wave dipole for use at 14 MHz, mounted at a height of λ/2 (10.7 m) above varying types of ground. The plot shows the radiation pattern in elevation, for a bearing of zero degrees in azimuth, where the radiation is a maximum, i.e. at right-angles to the line of the dipole. It can be seen that the size and shape of the main lobe are little affected by the ground conditions, whereas these strongly affect the radiation in the vertical direction, for the following reason.

Given the stated mounting height, the downward radiation reaches the ground in antiphase. Upon reflection, it suffers a phase reversal, so that the reflected wave at ground level is in phase with the upward radiation at the antenna itself. But by the time the reflected wave arrives back at the antenna, it is again in antiphase with the upward radiation at that point and therefore tends to cancel it. If the terrain beneath the antenna is a very good reflector, the reflected wave is barely reduced in amplitude, and so the cancellation is almost complete. Over poor ground, some of the energy radiated downwards penetrates the ground and is absorbed, whilst what is reflected may suffer a phase ‘reversal’ which is not exactly 180°. Thus the reflected wave arriving back at the antenna is reduced and cancellation is incomplete, leaving appreciable net radiation in the vertical direction.

If the antenna height is raised, the null (or minimum) in the vertical direction splits into two, either side of the vertical. The angular spacing between them increases as the height is raised further, with further nulls successively appearing and splitting likewise.

At a higher frequency, e.g. 30 MHz, the vertical radiation pattern of a horizontal halfwave dipole mounted at the same height in terms of λ, namely λ/2, is very similar to that of Figure 14.6. But mounted at the same physical height as the antenna of Figure 14.6, namely 10.7 m or approximately one wavelength, there will be two distinct lobes either side of the vertical. There is a deep null between them at an elevation angle of about 45°, where the radiation is 8 dB or more below isotropic in the case of good ground (high conductivity and permittivity) - much more in the case of sea water. On the other hand, with poor soil the null is only some 5 dB below isotropic, clearly better if the only path open to a distant receiver involves a take-off angle of 45°. Thus ‘good’ soil is not necessarily an advantage. With a 30 MHz half-wave horizontal dipole mounted at a height of 2λ m there are four lobes either side of the vertical. The deepest of these, at an elevation angle of about 14°, is very deep regardless of soil type, being some 10 dB below isotropic, the higher nulls being progressively less deep, except in the case of sea water. In many cases, HF communications are typically required over paths of a given length; mainly short paths - for example tactical comms - or alternatively mainly medium to long paths, e.g. diplomatic traffic. Thus an antenna mounting height would be chosen to avoid a null at the required take-off angle over the usual range of operating frequencies.

For a thin wire half-wave dipole, the aperture of 1.65λ2/4p square metres seems to bear little resemblance to the actual area, which is clearly much less than this. However, with a large antenna array, or a dish antenna, where the overall dimensions may be many wavelengths, it is found that the actual physical area does approximate to the effective area A = Gλ2/4p. For example, a microwave dish of physical area a, will have an effective area of A = 0.6a, approximately. (The factor 0.6 is due to the impossibility of designing a feed system which will distribute the power uniformly over the reflector without spilling any over the edge.) Thus at microwave where dish antennas are commonly employed, gains of 40-50 dB are available.

In all cases of directional antennas, the increased gain in the desired direction is bought at the expense of reduced gain in other directions. With high gain antennas, there are usually a number of ‘sidelobes’, directions in which the gain, though much less than that in the main lobe, is nevertheless considerable. In some cases a directional antenna is employed more to discriminate against unwanted signals coming from a different direction from the wanted signal, than to increase the gain to the latter. A common example is in TV reception, where an antenna with a high front-to-back ratio can reduce ghosting due to reflections of the wanted signal, or interference due to another station. The examples just given are mostly terrestrial situations; only in space applications or in microwave links using very directional dish antennas will the free-space path loss formula be applicable.

So far, only individual antennas have been considered. The chapter would not be complete, however, without some mention of antenna arrays. These may be used for a number of purposes. For instance, an in-line array of antennas, all fed with equal amounts of power in the same phase from a transmitter via a splitter, will produce narrow beams, like a long thin figure-of-eight at right angles to the array, plus various sidelobes. On the other hand, if each individual antenna is fed with the signal, in equal amounts but suitably successively delayed in phase, a narrow end-fire beam is effected. Such linear arrays, given the necessary adjustable phasing arrangements, can be used as directional receiving antenna systems, and hence also as DF (direction finding) systems. Circular arrays of monopoles are used in DF systems at HF, such as in the Wullenweber system (where the large aperture permits the synthesis of narrow beams, especially in the upper part of the HF band) and at VHF, e.g. short range coastal DF installations. Compact arrays are necessary where space is limited, e.g. the Bellini-Tosi antenna (consisting of crossed triangular loops connected to a goniometer) once commonly used for ship-borne DF. Another example is the Adcock DF antenna (consisting of four vertical half-wave dipoles mounted at the ends elevated cross-arms and connected to phase-difference measuring equipment) for tactical DF applications, where rapid redeployment is a requirement.

A major accuracy limitation in DF systems, at both HF and VHF, is due to the reception of different rays, i.e. different versions of the same signal via different paths. At HF, these will usually be different skywave paths, whilst at VHF there may be both direct and reflected rays; both are examples of multipath propagation. In addition, in the tactical environment there is often great interest in DF on co-channel signals.

The simplest fixed two antenna array can give the bearing of an intercepted signal, but cannot distinguish between the true bearing and a ‘reciprocal’ bearing, at 180°. In some applications, e.g. in a tactical military environment, this will not be a problem, since enemy signals will originate from beyond the FLOT (front line of own troops). In other cases, the ambiguity can be resolved if the antenna array can be rotated slightly - if the array is moved anticlockwise, the phase of the signal in the right-hand element will advance relative to that in the left-hand element provided the signal source is in front of the array, or retard if behind. This the basis of some covert vehicle-borne tracking systems.

If the array is receiving one unique signal, via a single path, the amplitude of the signals from the two antennas will be equal, at least when the target is dead ahead. If they differ, this indicates that the signal is comprised of two ‘rays’ or wavefronts, arriving from slightly different directions. On the basis of an assumption that both arrive at the same low elevation angle, likely at VHF but unlikely at HF, it may be possible to estimate the two rays, but in many cases, e.g. with HF signals or more than two rays, this will be impossible. If an additional antenna or two is added to the array, much more data become available, and the process has been carried further.

Figure 14.14 is the block diagram of the hardware developed to run advanced SRDF (super-resolution direction finding) algorithms such as MUSIC (multiple signal classification) on signals gathered from an eight antenna array. The system copes with multi-path reception of signals and with multiple signals on the same channel. The SRDF algorithms require knowledge of the antenna array layout, i.e. the relative x, y and z (height) co-ordinates of the individual antennas, some irregularity in the layout being positively beneficial. Using SRDF, small tactical arrays can provide similar performance to that previously only achieved with much larger fixed site arrays [9]. The system also provides an adaptive beam-forming capability. This permits a beam to be formed in the direction of an intercepted signal of interest, whilst simultaneously steering nulls in the direction of all other signals, subject to there not being too many relative to the number of antennas in the array.

Many member states of the ITU maintain monitoring stations, to help police the usage of frequency allocations etc. The UK Radio Agency’s monitoring station at Baldock now uses an SRDF system, making use of a subset of antennas in a pre-existing large HF receiving array, and similar systems are in use in North America, Australia and various European countries.

SRDF systems can also provide good results in difficult installations, such as on board warships. Here, the most favoured antenna position, at the top of the highest mast, is generally not available to a DF installation, and all other antenna locations naturally suffer severe local multi-path, due to reflections from the ship’s superstructure. Special techniques, including calibration against known targets over the full range of bearings in azimuth, enable SRDF algorithms to work under these extreme conditions.

Finally, there are specialized antennas for field-strength measurements. These are covered in Chapter 16, Measurements.