Chapter 16

Power Supplies

Power for Phono Preamplifiers

Moving-magnet cartridges generate small signals, at about 5 mV rms, while moving-coil cartridges give even smaller ones, down to 50 uV in some cases. This might lead you to suppose that very clean power supplies are essential to get the best noise and hum figures, but as with so many things that seem obvious in audio, it is not the case. I have designed many phono preamps that were powered directly by 78/79 or LM317/337 type IC regulators, with no additional filtering. In no case did rail noise or ripple make any detectable contribution to the output. To give this some perspective, professional microphone preamplifiers work with signal down to −80 dBu, which is 77 uV rms, and I have powered thousands of them directly from supply rails provide by quite conventional power supplies, with no hint of a problem. They were of course balanced preamplifiers, and this undoubtedly helps.

This immunity rail noise or ripple is very largely due to the excellent power supply rejection ratio (PSRR) of opamps such as the 5532 or the LM4562, but there are some important points to be made.

1. 1) The decoupling capacitor grounds must be taken back to the point where power enters the PCB to prevent them injecting noise or ripple into the signal ground. In particular the input ground and the ground at the bottom of the negative-feedback network must be at the same potential, and allowing decoupling currents to flow through this sensitive part of the circuit will be disastrous.
2. 2) It is assumed earlier that the phono preamp consists only of opamps with high PSRR. Some configurations however use a discrete transistor or transistors as a front end to an opamp, to reduce noise, and these must be powered from one or both supply rails. Single- transistor configurations in particular are likely to have no PSRR at all, and heavy filtering of the supply will be necessary; simple R-C filters are usually all that is required, but the capacitances can get quite large (say 1000 uF) if the voltage drop across the associated resistor is to be kept low. Configurations with two discrete devices in a differential pair are likely to show some PSRR, but it will not be of the same order as an opamp, and supply filtering will almost certainly still be necessary.
3. 3) It is assumed that the power supply is properly designed so that the IC regulators give the performance they are capable of. Specifically, ripple currents on the unregulated side must be kept out of the reference and regulated output circuitry. This is not hard to do.
4. 4) The most difficult cause of power-supply-related hum problems is not so much the supply rails as the magnetic field from the mains transformer. If you have this inside the same box as the phono preamp, then in my experience it will define the hum performance of the circuitry. A separate external supply, even if it is only a humble wall-wart, i.e. plug-type AC adapter, gives physical distance and is the only real solution to this issue; there is more on this at the end of the chapter.

Opamp Supply Rail Voltages

It has been mentioned several times in the earlier chapters of this book that running opamps at the slightly higher voltage of ±17V rather than ±15V gives an increase in headroom and dynamic range of 1.1 dB for virtually no cost and with no reliability penalty. Soundcraft ran all the opamps in their mixing consoles at ±17V for at least two decades, and opamp failures were almost unknown. This recommendation assumes that the opamps concerned have a maximum supply voltage rating of ±18V, which is the case for the Texas TL072, the new LM4562, and many other types.

The 5532 is (as usual) in a class of its own. Both the Texas and Fairchild versions of the NE5532 have an absolute maximum power supply voltage rating of ±22V, (though Texas also gives a “recommended supply voltage” of ±15V), but I have never met any attempt to make use of this capability. The 5532 runs pretty warm on ±17V when it is simply quiescent, and my view (and that of almost all the designers I have spoken to) is that running it at any higher voltage is simply asking for trouble. This is a particular concern in the design of mixing consoles, which may contain thousands of opamps—anything that that impairs their reliability is going to cause a lot of trouble. In any case, moving from ±17V rails to ±18V rails only gives 0.5 dB more headroom. Stretching things to ±20V would give 1.4 dB more than ±17V, and running on the ragged edge at ±22V would yield a more significant 2.2 dB more than ±17V, but you really wouldn’t want to do it. Pushing the envelope like this is also going to cause difficulties if you want to run opamps with maximum supply ratings of ±18V from the same power supply.

We will therefore concentrate here on ±17V supplies for opamps, dealing first with what might be called “small power supplies” i.e. those that can be conveniently built with TO-220 regulators. This usually means an output current capability that does not exceed 1.5 amps, which is plenty for even complicated phono amplifiers, preamplifiers etc.

An important question is: how low does the noise and ripple on the supply output rails need to be? Opamps in general have very good power supply rejection ratios (PSRR), and some manufacturer’s specs are given in Table 16.1.

The PSRR performance is actually rather more complex than the bare figures given in the table imply; PSRR is typically frequency-dependent (deteriorating as frequency rises) and different for the +V and −V supply pins. It is however rarely necessary to get involved in this degree of detail. Fortunately even the cheapest IC regulators (such as the venerable 78xx/79xx series) have low enough noise and ripple outputs that opamp PSRR performance is rarely an issue.

Table 16.1 PSRR specs for common opamps

Opamp type	PSRR minimum dB	PSRR typical dB
5532	80	100
LM4562	110	120
TL072	70	100

There is however another point to ponder; if you have a number of electrolytic-sized decoupling capacitors between rail and ground, enough noise and ripple can be coupled into the nonzero ground resistance to degrade the noise floor. Intelligent placing of the decouplers can help—putting them near where the ground and supply rails come onto the PCB means that ripple will go straight back to the power supply without flowing through the ground tracks on the rest of the PCB. This is of limited effectiveness if you have a number of PCBs connected to the same IDC cable, as in many small mixing desks, and in such cases low ripple power supplies may be essential.

Apart from the opamp supply rails, audio electronics may require additional supplies, as shown in Table 16.2.

It is often convenient to power relays from a +9V unregulated supply that also feeds the +5V microcontroller regulator—see later in this chapter. The use of +24V to power LED metering systems is dealt with in Chapter 15 on metering.

Designing a ±15V Supply

Making a straightforward ±15V 1 amp supply for an opamp-based system is very simple, and has been ever since the LM7815/7915 IC regulators were introduced (which was a long time ago). They are robust and inexpensive parts with both overcurrent and over- temperature protection and give low enough output noise for most purposes. We will look quickly at the basic circuit because it brings out a few design points which apply equally to more complex variations on the theme. Figure 16.1 shows the schematic, with typical component values; a centre-tapped transformer, a bridge rectifier, and two reservoir capacitors C1, C2 provide the unregulated rails that feed the IC regulators. The secondary fuses must be of the slow-blow type. The small capacitors C7–C9 across the input to the bridge reduce RF emissions from the rectifier diodes; they are shown as X-cap types not because they have to withstand 230 Vrms but to underline the need for them to be rated to withstand continuous AC stress. The capacitors C3, C4 are to ensure HF stability of the regulators, which like a low AC impedance at their input pins, but these are only required if the reservoir capacitors are not adjacent to the regulators, i.e. more than 10 cm away. C5, C6 are not required for regulator stability with the 78/79 series—they are there simply to reduce the supply output impedance at high audio frequencies.

Table 16.2 Typical additional supply rails for opamp-based systems

Supply voltage	Function
+5V	Housekeeping microcontroller
+9V	Relays
+24V	LED bargraph metering systems, discrete audio circuitry, relays

There are really only two electrical design decisions to be made; the AC voltage of the transformer secondary and the size of the reservoir capacitors. As to the first, you must make sure that the unregulated supply is high enough to prevent the rails dropping out (i.e. letting hum through) when a low mains voltage is encountered but not so high that either the maximum input voltage of the regulator is exceeded or it suffers excessive heat dissipation. How low a mains voltage it is prudent to cater for depends somewhat on where you think your equipment is going to be used, as some parts of the world are more subject to brown-outs than others. You must consider both the minimum voltage drop across the regulators (typically 2V) and the ripple amplitude on the reservoirs, as it is in the ripple troughs that the regulator will first “drop out” and let through unpleasantness at 100 Hz.

In general, the RMS value of the transformer secondary will be roughly equal to the DC output voltage.

The size of reservoir capacitor required depends on the amount of current that will be drawn from the supply. The peak-to-peak ripple amplitude is normally in the region of 1 to 2 volts; more ripple than this reduces efficiency because the unregulated voltage has to be increased to allow for unduly low ripple troughs, and less ripple is usually unnecessary and gives excessive reservoir capacitor size and cost. The amount of ripple can be estimated with adequate accuracy by using Equation 16.1

$Vpkpk=\frac{I\cdot \Delta t\cdot 1000}{C}$

16.1

where:

Vpk-pk is the peak-to-peak ripple voltage on the reservoir capacitor.

I is the maximum current drawn from that supply rail in amps.

Δt is the length of the capacitor discharge time, taken as 7 milliseconds.

C is the size of the reservoir capacitor in microfarads.

The “1000” factor simply gets the decimal point in the right place.

Note that the discharge time is strictly a rough estimate and assumes that the reservoir is being charged via the bridge for 3 msec and then discharged by the load for 7 msec. Rough estimate it may be, but I have always found it works very well.

The regulators must be given adequate heatsinking. The maximum voltage drop across each regulator (assuming 10% high mains) is multiplied by the maximum output current to get the regulator dissipation in watts, and a heatsink selected with a suitable thermal resistance to ambient (in °C per watt) to ensure that the regulator package temperature does not exceed, say, 90 °C. Remember to include the temperature drop across the thermal washer between regulator and heatsink.

Under some circumstances it is wise to add protective diodes to the regulator circuitry, as shown in Figure 16.2. The diodes D1, D3 across the regulators are reverse-biased in normal operation, but if the power supply is driving a load with a large amount of capacitance, it is possible for the output to remain higher in voltage than the regulator input as the reservoir voltage decays. D1, D3 prevent this effect from putting a reverse voltage across the regulators. Such diodes are not usually required with normal opamp circuitry, as the amount of rail decoupling, shown as C7, C8 in Figure 16.2, is usually modest.

The shunt protection diodes D2, D4 are once again reverse-biased in normal operation. D2 prevents the +15V supply rail from being dragged below 0V if the −15V rail starts up slightly faster, and likewise D4 protects the −15V regulator from having its output pulled above 0V. This can be an important issue if rail-to-rail decoupling such as C9 is in use; such decoupling can be useful because it establishes a low AC impedance across the supply rails without coupling supply rail noise into the ground, as C7, C8 are prone to do. However, it also makes a low-impedance connection between the two regulators. D2, D4 will prevent damage in this case but leave the power supply vulnerable to start-up problems; if its output is being pulled down by the −15V regulator, the +15V regulator may refuse to start. This is actually a very dangerous situation, because it is quite easy to come up with a circuit where start-up will only fail one time in twenty or more, the incidence being apparently completely random but presumably controlled by the exact point in the AC mains cycle where the supply is switched on and other variables such as temperature, the residual charge left on the reservoir capacitors, and the phase of the moon. Do not fit C9.

If even one start-up failure event is overlooked or dismissed as unimportant, then there is likely to be serious grief further down the line. Every power supply start-up failure must be taken seriously.

Designing a ±17V Supply

There are 15 V IC regulators (7815, 7915) and there are 18 V IC regulators (7818, 7918), but there are no 17 V IC regulators. This problem can be effectively solved by using 15 V regulators and adding 2 volts to their output by manipulating the voltage at the REF pin. The simplest way to do this is with a pair of resistors that divide down the regulated output voltage and apply it to the REF pin, as shown in Figure 16.3a. (The transformer and AC input components have been omitted in this and the following diagrams, except where they differ from those shown earlier). Since the regulator maintains 15V between the OUT and REF pin, with suitable resistor values the actual output with respect to 0V is 17V.

The snag with this arrangement is that the quiescent current that flows out of the REF pin to ground is not well controlled; it can vary between 5 and 8 mA, depending on both the input voltage and the device temperature. This means that R1 and R2 have to be fairly low in value so that this variable current does not cause excessive variation of the output voltage, and therefore power is wasted.

If a transistor is added to the circuit as in Figure 16.3b, then the impedance seen by the REF pin is much lower. This means that the values of R1 and R2 can be increased by an order of magnitude, reducing the waste of regulator output current and reducing the heat liberated. This sort of manoeuvre is also very useful if you find that you have a hundred thousand 15 V regulators in store, but what you actually need for the next project is an 18 V regulator, of which you have none.

What about the output ripple with this approach? I have just measured a power supply using the exact circuit of Figure 16.3b, with 2200 uF reservoirs, and I found −79 dBu (87 uV rms) on the +17 V output rail and −74 dBu (155 uV rms) on the 17 V rail, which is satisfyingly low for inexpensive regulators and should be adequate for almost all purposes; note that these figures include regulator noise as well as ripple. The load current was 110 mA. If you are plagued by ripple troubles, the usual reason is a rail decoupling capacitor that is belying its name by coupling rail ripple into a sensitive part of the ground system, and the cure is to correct the grounding rather than design an expensive ultra-low ripple PSU. Note that doubling the reservoir capacitance to 4400 uF only improved the figures to −80 dBu and −76 dBu respectively; just increasing reservoir size is not a cost-effective way to reduce the output ripple.

Using Variable-Voltage Regulators

It is of course also possible to make a ±17V supply by using variable output voltage IC regulators such as the LM317/337. These maintain a small voltage (usually 1.2V) between the OUTPUT and ADJ (shown in figures as GND) pins and are used with a resistor divider to set the output voltage. The quiescent current flowing out of the ADJ pin is a couple of orders of magnitude lower than for the 78/79 series, at around 55 uA, and so a simple resistor divider gives adequate accuracy of the output voltage, and transistors are no longer needed to absorb the quiescent current. A disadvantage is that this more sophisticated kind of regulator is somewhat more expensive than the 78/79 series; at the time of writing they cost something like 50% more. The 78/79 series with transistor voltage-setting remains the most cost-effective way to make a non-standard-voltage power supply at the time of writing.

It is clear from Figure 16.4 that the 1.2 V reference voltage between ADJ and out is amplified by many times in the process of making a 17 V or 18 V supply; this not only increases output ripple but also output noise, as the noise from the internal reference is being amplified. The noise and ripple can be considerably reduced by putting a capacitor C7 between the ADJ pin and ground. This makes a dramatic difference; in a test PSU with a 650 mA load, the output noise and ripple was reduced from −63 dBu (worse than 78xx series) to −86 dBu (better than 78xx series), and so such a capacitor is usually fitted as standard. If it is fitted, it is then essential to add a protective diode D1 to discharge C7, C8 safely if the output is short-circuited, as shown in Figure 16.5.

The ripple performance of the aforementioned test PSU, with a 6800 uF reservoir capacitor and a 650 mA load, is summarised for both types of regulator in Table 16.3. Note that the exact ripple figures are subject to some variation between regulator specimens.

Improving Ripple Performance

Table 16.3 shows that the best noise and ripple performance that can be expected from a simple LM317 regulator circuit is about −86 dBu (39 uV rms), and this still contains a substantial ripple component. The reservoir capacitors are already quite large at 4700 uF, so what is to be done if lower ripple levels are needed? The options are:

1. 1) Look for a higher-performance IC regulator. They will cost more, and there are likely to be issues with single sourcing.
2. 2) Design your own high-performance regulator using discrete transistors or opamps. This is not a straightforward business if all the protection that IC regulators have is to be included. There can also be distressing issues with HF stability.
3. 3) Add an RC input filter between the reservoir capacitor and the regulator. This is simple and pretty much bullet-proof and preserves all the protection features of the IC regulator, though the extra components are a bit bulky and not that cheap. There is some loss of efficiency due to the voltage drop across the series resistor; this has to be kept low and the capacitance large.

Table 16.3 Comparing the noise and ripple output of various regulator options

	7815 + transistor dBu	LM317 dBu	LM317 uV
No C on LM317 ADJ pin	−73 dBu (all ripple)	−63 dBu (ripple & noise)	549 uV
47 uF on LM317 ADJ pin	−73 dBu (all ripple)	−86 dBu (ripple & noise)	39 uV
Input filter 2.2Ω & 2200 uF	−78 dBu (ripple & noise)	−89 dBu (mostly noise)	27 uV
Input filter 2.2Ω & 4400 uF	−79 dBu (mostly noise)	−90 dBu (all noise)	24 uV

The lower two rows of Table 16.3 show what happens. In the first case the filter values were 2.2 Ω and 2200 uF. This has a −3 dB frequency of 33 Hz and attenuates the 100 Hz ripple component by 10 dB. This has a fairly dramatic effect on the output ripple, but the dB figures do not change that much as the input filter does not affect the noise generated inside the regulator. Increasing the filter capacitance to 4400 uF sinks the ripple below the noise level for both types of regulator.

Dual Supplies From a Single Winding

It is extremely convenient to use third-party “wall-wart” power supplies for small pieces of equipment, as they come with all the safety and EMC approvals already done for you, though admittedly they do not look appropriate with high-end equipment. The problem is that the vast majority of these supplies give a single AC voltage on a two-pole connector, so a little thought is required to derive two supply rails. Figure 16.6 shows how it is done in a ±18V power supply; note that these voltages are suitable only for a system that uses 5532s throughout. Two voltage-doublers of opposite polarity are used to generate the two unregulated voltages. When the incoming voltage goes negative, D3 conducts and the positive end of C1 takes up approximately 0V. When the incoming voltage swings positive, D1 conducts instead and the charge on C1 is transferred to C3. Thus the whole peak-to-peak voltage of the AC supply appears across reservoir capacitor C3. In the same way, the peak- to-peak voltage, but with the opposite polarity, appears across reservoir C4.

Since voltage-doublers use half-wave rectification, they are not suitable for high current supplies, but this arrangement should be more than adequate for most phono preamplifiers. When choosing the value of the reservoir capacitor values, bear in mind that the discharge time in Equation 16.1 must be changed from 7 msec to 17 msec. The input capacitors C1, C2 should be the same size as the reservoirs.

Power supplies for discrete circuitry

One of the main reasons for using discrete audio electronics is the possibility of handling larger signals than can be coped with by opamps running off ±17 V rails. The use of ±24 V rails allows a 3 dB increase in headroom, which is probably about the minimum that justifies the extra complications of discrete circuitry. A ±24 V supply can be easily implemented with 7824/7924 IC regulators. On the other hand you have to consider that stages downstream may have opamps running off ±17 V rails, and you don’t want to supply signal voltages that will blow them up. Hi-fi equipment rarely if ever has over-voltage protection.

A slightly different approach was used in my first published preamplifier design.[1] This preamp in fact used two LM7824 +24V regulators connected as shown in Figure 16.7 because, at the time, the LM7924–24V regulator had not yet reached the market. The use of a second positive regulator to produce the negative output rail looks a little strange at first sight, but I can promise you it works. It can be very useful in the sort of situation described earlier; you have a hundred thousand +15V regulators in store but no −15V regulators … I’m sure you see the point.

Note that this configuration requires two completely separate transformer windings; it cannot be used with a centre-tapped secondary.

Mutual Shutdown Circuitry

It is an awkward quirk of 5532 opamps that if one supply rail is lost and collapses to 0V, while the other rail remains at the normal voltage, they can under some circumstances get into an anomalous mode of operation that draws large supply currents and ultimately destroys the opamp by over-heating. To prevent damage from this cause, which could be devastating to a large mixing console, the opamp supplies are very often fitted with a mutual shutdown system. Mutual shutdown ensures that if one supply rail collapses because of overcurrent, over-temperature, or any other cause, the other rail will be promptly switched off. The extra circuitry required to implement this is shown in Figure 16.8, which is an example of a high current supply using 7.5 amp regulators.

The extra circuitry to implement mutual shutdown in Figure 16.8 is very simple; R5, D3, R6 and Q1 and Q2. Because R5 is equal to R6, D3 normally sits at around 0 V in normal operation. If the +17V rail collapses, Q2 is turned on by R6, and the REF pin of U2 is pulled down to the bottom rail, reducing the output to the reference voltage (1.25 V). This is not completely off, but it is low enough to prevent any damage to opamps.

If the −17 V rail collapses, Q1 is turned on by R5, pulling down the REF pin of U1 in the same way. Q1 and Q2 do not operate exactly symmetrically, but it is close enough for our purposes.

Note that this circuit can only be used with variable output voltage regulators because it relies on their low reference voltages.

Microcontroller and Relay Supplies

It is very often most economical to power relays from an unregulated supply. This is perfectly practical, as relays have a wide operating voltage range. If 9 V relays are used, then the same unregulated supply can feed a +5 V regulator to power a microcontroller, as shown in Figure 16.9.

Hum induced by electrostatic coupling from an unregulated relay supply rail can be sufficient to compromise the noise floor; the likelihood of this depends on the physical layout, but inevitably the signal paths and the relay supply come into proximity at the relay itself. It is therefore necessary to give this rail some degree of smoothing, without going to the expense of another regulator and heatsink. (There must be no possibility of coupling between signal ground and relay power ground; these must only join right back at the power supply.) This method of powering relays is more efficient than a regulated rail because it does not require a voltage drop across a regulator that must be sufficient to prevent drop-out and consequent rail ripple at low mains voltages.

Simple RC smoothing works perfectly well for this purpose. Relays draw relatively high currents, so a low R and a high-value C are used to minimise voltage losses in R and changes in the relay supply voltage as different numbers of relays are energised.

The RC smoothing values shown in Figure 16.9 are typical but are likely to need adjustment depending on how many relays are powered and how much current they draw. R1 is low at 2.2Ω and C2 high at 4700 uF; fortunately the voltage is low, so C2 need not be physically large.

Mains Transformers and Magnetic Fields

If you are constrained to put the mains transformers in the same box as the phono preamplifier, there are several points to consider:

1. 1) Transformer type.
    In these times the transformer will almost certainly be a toroid. I would resist the temptation to save a bit on money by using an open-frame transformer, as they have larger external magnetic fields. Open-frame transformers should only be considered for external power supplies.
2. 2) Transformer position.
    Put the transformer as far away as possible from the sensitive circuitry. Hopefully you will be able to put the phono preamplifier at one end of the box and the transformer at the other.
3. 3) Transformer orientation.
    It is common practice to rotate a toroid about its central bolt to minimise induced hum. It is not usually economic to optimise the toroid orientation for each example of a product, but toroids made by reputable manufacturers should not vary much in the shape of their hum field, and the orientation can be fixed at the design stage. Unfortunately it’s not quite so simple as just “turn for zero hum”. There may be an obvious minimum, but it is rather unlikely that you will get zero hum; the transformer field is unlikely to be getting into the electronics at just one point.
    If the susceptible electronics is spread out over space, perhaps with left and right channels on opposite sides of the enclosure (not a good idea for preamplifiers), then with dreadful certainty it will be found that the hum minimum for one channel is something like the maximum for the other. Even if the two channels are immediately adjacent, you are unlikely to be able to get a good minimum on both at once.
    Some toroids have single-strand secondary lead-outs, which are too stiff to allow rotation; for experimental toroid-turning these can be cut short and connected to suitably large flexible wire such as 32/02, with carefully sleeved and insulated joints.
4. 4) Transformer screening.
    The external field of a toroidal transformer can be reduced by wrapping the outside of the toroid in one or more layers of silicon steel, the intention being screening rather than the creation of a shorted turn. The success of this depends on using high-quality silicon steel, or better still GOSS (grain-oriented silicon steel), and even then the reduction in hum figures from the affected circuitry is not likely to be more than 6 dB. It may sound unlikely, but it is a fact that the method of making GOSS was discovered in 1935—by a Mr N. P. Goss. Mumetal, a nickel-iron alloy (75% nickel, 15% iron, plus copper and molybdenum) is an even more effective magnetic screening material, but it is expensive and has a disconcerting habit of losing its magical properties if bent or otherwise deformed, and according to some authorities if you just drop it on the floor.
5. 5) Transformer manufacture.
    There are transformer manufacturers with a reputation for making low-field transformers. At least one toroid manufacturer specialises in low-field designs for audio applications, and their products can be have an external field 10 dB lower than a standard-quality toroid transformer. On the downside, the price will be something like twice as much. Low-field transformers usually incorporate GOSS screening as described earlier, but there are also changes to the internal flux levels and so on, so the toroids are usually slightly larger than a conventional design.
6. 6) Extra shielding.
    In dire circumstances you might consider adding extra walls of GOSS or mumetal between the sensitive circuitry and the transformer. I have seen this done, but it was expensive and not very effective.

From these statements you can see that the options available for reducing transformer hum are strictly limited. The transformer field might be called the transmitting end of the problem, while the receiving end is the electronics of the phono preamplifier, and it is obviously helpful to minimise its sensitivity to magnetic fields. The basic principle is simple—make sure that the loop area between signal conductors and the signal ground is as small as possible. This is relatively easy for input connections, where it is of course crucial, but much harder to do on a PCB full of components, and with NFB paths and so on. I have spent time cutting tracks and replacing them with bits of solid-core that could be rerouted, and my conclusion is that it is a thankless task and unlikely to give much improvement. Every new layout requires a PCB iteration to test it properly, and that takes time and money.

External Power Supplies

Even if you follow all the precautions in the previous section, it is very difficult to keep all traces of transformer-induced hum out of the signal circuitry. It is highly irritating to find that despite the cunning use of low-noise circuitry, the noise floor is defined by the deficiencies of a component—for the ideal transformer would obviously have no external field—rather than the laws of physics as articulated by Johnson. The authoritative solution is to put the mains transformer in a separate box which can be placed a metre or more away from the preamplifier unit, which is powered through an umbilical lead. A cable that is captive at the power supply end saves the cost of a mating pair of connectors, which may be considerable. The supply voltages involved will probably be below the 50 volt limit set by the Low Voltage Directive, so there is no need to take elaborate precautions to ensure that the connector contacts cannot be touched.

Advantages:

• The transformer field hum problem is completely solved.
• It will appeal to some potential customers as a “serious” approach to high-end audio.
• There are no high-voltage AC mains inside the preamp box. If you are going to encourage the customer to open the box to change input loading capacitance, etc., this is essential for safety reasons. In general you don’t want customers opening boxes; in the world of safety regulations, if you need a tool such as a screwdriver to open the box, that should give you a hint you shouldn’t be doing it.

Disadvantages:

• The cost of an extra enclosure plus an extra cable and connectors, power indicator lights etc. The connectors will probably have to be multi-pole. The transformer box must have fuses or other means of protection in case of short-circuits in the cable.
• In my experience a significant proportion of users will, exhortations to the contrary notwithstanding, promptly place the amplifier box directly on top of the transformer box, immediately defeating the whole object. This is particularly likely if the two boxes have the same footprint and so look as if they ought to be stacked together. However, all may not be lost in this situation, as the transformer is still physically further away from the sensitive electronics (though if the transformer has a large field emerging from its ends things may actually be worse), and there are now two extra layers of steel interposed (assuming the boxes are made of steel, that is). Unfortunately plain sheet steel is not good at stopping magnetic fields and may even increase the coupling.

The cheapest and simplest external supply is a plug-type AC adapter, i.e. a wall-wart. Higher-power versions are usually fitted in the cable and are called “line lumps”. The adapter produces low voltage AC either with a conventional transformer or a small switch-mode supply. The latter should be avoided if you want power free from high-frequency interference. The output is isolated from the mains, and to underline the point, the ground pin (I’m thinking here of UK 13 amp plugs) is usually plastic, and its only function is to lift the safety shutter over the live and neutral contacts so the plug can be inserted. These adapters come with safety and EMC approvals done for you, with the important proviso that you choose a reputable supplier. There are dodgy versions available that may be covered with fake safety approvals and CE marks but are actually a serious safety hazard; please take this issue seriously. Fraudulent marking of substandard adapters with the name of reputable manufacturers is known,[2] and a trustworthy supplier is essential.

Most AC adapters produce a single voltage on a two-pin connector; generating + and − supply rails from this is no problem so long as you only need a small amount of power; see the earlier section “Dual Supplies From a Single Winding”. Centre-tap outputs on three-pin connectors are available but noticeably more expensive due to much lower production levels.

The use of an AC adapter naturally means that mains-frequency power is brought inside the preamp box, and rectification, smoothing, and regulation take place in there too. While this is perhaps not ideal, the problems are tiny compared with the magnetic field of a transformer, and there is no reason why the audio performance should be in any way compromised given suitable care in design. There is the disadvantage that the current in the umbilical cable will consist of short but relatively large charging pulses at 100 or 120 Hz, so the cable needs to kept away from audio leads. The pulses give a greater voltage drop in the cable resistance than a steady current but also give rise to much greater I²R heating. At these low power levels the latter is unlikely to cause problems in the cable itself but can be fatal to the contacts of connectors. Speaking from bitter experience, I can warn you that connectors that appear to have a more than adequate safety margin can fail under these conditions, so be cautious with ratings and do plenty of testing.

If you are building your own power supply from scratch in its separate box, there are more options:

1. 1) Put not just the mains transformer but also the rectifiers and reservoir capacitors in the power supply box. The current in the umbilical cable is now rectified and smoothed DC, which is much less likely to interfere with anything, and it is much easier to specify connectors to cope with it. The regulators are placed in the preamp box, so their outputs can be directly connected to the audio electronics, and the impedance of the DC supply cable is not an issue. Depending on the design of the connectors, it may be necessary to provide short circuit protection for the smoothed DC, either with plain fuses or, better, with positive-tempco polyfuses that self-reset.
2. 2) Put the mains transformer, rectifiers, reservoir capacitors and regulators in the power supply box. The current in the umbilical cable is now regulated DC, which cannot interfere with anything, and there are no connector issues. The downside is that the supply cable impedance is now in series with the supply to the audio circuitry. Some sizable decoupling capacitors may be needed at the preamp end. Short circuit protection will be provided by the regulators.

My final thought here is that I would rather use a cheap Chinese wall-wart with a metre-long lead than have the most super-expensive toroid in the world inside the preamp box.

References

 1. Self, D. “An Advanced Preamplifier Design” Wireless World, Nov 1976.

 2. https//en.wikipedia.org/wiki/AC_adapter Accessed Sept 2016.