CMOS THz Imaging 267
Figure 12.7: Absorption ratio of various types of oil detected at 135
GHz.
12.3 240 ∼ 280-GHz Wide-Band Imager with
Heterodyne Receiver
Since THz radiation is highly sensitive to the crystal lattice vibration,
hydrogen-bond as well as intermolecular interactions, it results in unique spec -
troscopy finger prints for many materials. There are two design targets for the
receiver to enable the spectroscopy analysis in a sub-THz imaging: high spec-
trum selectivity and wide frequency range of operation. Diode detection-based
receivers [230, 32, 259] can achieve the latter target, but not the former one;
while super-rege ne rative-based receivers [90, 231] can achieve the former tar-
get, but no t the latter one. In order to satisfy both of the two targets in
the receiver design, one needs to deploy either heterodyne [264] or direct-
conversion receivers. Compared to direct-conversion architecture with a zero
IF, heterodyne architecture with a near-zero IF is more flexible in the sys tem
design with both magnitude and phase detection capability. The magnitude
detection in an s ub-THz imaging system can be achieved by either heterodyne
or direct-conversion architecture, but only heterodyne architecture is able to
retain the phase information of sub-THz signal, which is very useful to ana-
lyze the c omplex refractive index of the sample under test. As such, this paper
fo cuses on the design of heterodyne receivers with a near-zero IF.
268 Design of CMOS Millimeter-Wave and Terahertz Integrated Circuits
12.3.1 Architecture and System Specification
The de sign of a CMOS heterodyne receiver in THz has to be conducted in a
scenario without any low noise amplifiers (LNA) as illustrated in Figure 12.2,
because hardly any amplifiers can be designed at a frequency close to or above
the f
max
. For example, the f
max
in a typical CMOS 65nm process is around
300 GHz [26 5],
The receiver gain (G
tot
) in such c ase can be calculated as
G
tot
= G
ant
G
mix
G
pga
(12.4)
where G
ant
, G
mix
and G
pga
denote the gain of antenna, mixer and power gain
amplifier (PGA), respectively. Also, the total rece iver noise figure (NF
tot
)
becomes
NF
tot
= NF
mix
+
NF
pga
− 1
G
mix
(12.5)
where NF
mix
and NF
pga
denote the NF of the mixer and VGA, respectively.
Eq. (12.4) denotes that the total receiver gain can be improved from antenna,
mixer and PGA, while (12.5) deno tes that the noise contributed by each sta ge
decreases as the total gain of preceding stages. The noise contributions from
mixer and PGA are no longer negligible witho ut LNA, so the NF of the
receiver has to be improved by increas ing the conversion gain of the mixer
and minimizing the noise fig ure of PGA. In the following sections, the designs
of the mixer and PGA in a THz CMOS heterodyne receiver are introduced.
12.3.2 Down-Conversion Mixer
As the first active building block connected to the antenna, the design of the
down-conversion mixer largely affects the performance of heterodyne receiver,
including conversio n gain and NF. Co nventionally there are two types of mix-
ers that are commonly used in the mm-wave region: Gilbert-cell mixer and
single-ga te mixer [266, 267]. Gilbert-cell mixer [266] has a compact size and
low implementation loss , and it generates the cross-modulation product of LO
and RF signals. However, the co nversion gain of the Gilbert-cell mixer largely
depe nds on the transconductance of transistors in the saturation re gion, which
will be heavily reduced when the signal freque nc y is approaching f
max
. On the
other hand, the single-gate mixer [267] utilized the nonlinea rity of transistors
when biased in the subthreshold region, which is less frequency dependent
compared to the Gilbert-cell mixer, and is able to work at a higher frequency
in THz. Moreover, compared to the subharmonic mixer working with 1/3-LO
[234], the conversion loss could be larg ely reduced when directly mixing the
RF and LO signal in fundamental tones. In this work, a single-gate mixe r is
designed to down-convert the RF signal in 220 ∼ 300 GHz to the baseband
by the fundamental tone of LO signal.
Figure 12.8 shows the schematic of a proposed down-conversion mixer
design. A Wilkinson combiner implemented by coplanar waveguide (CPW)
CMOS THz Imaging 269
Figure 1 2.8: Schematic of THz d own-conversion mixer at 280 GHz.
is deployed to combine the RF signal from the antenna and the LO signal,
and also to provide isolation in between. The combined output is connected
to the input of the common-source stage (M1) biased in the subthreshold
region (0.4-V V
GS
) by a compact compos ite CPW and lump compo ne nts
matching network. The following c ommon-gate stage is applied to improve
the conversion efficiency a s well as the reverse isolation. One LC resonator is
connected between the VDD and the mixer output to filter out the unwanted
harmonics of the IF signal.
A post-layout simulation is performed to the proposed mixer design with
passive devices s imulated in an EMX environment, including power combiner,
matching network and inductors. A maximum conversion gain of -19 dB is
demonstrated by proposed mixer in Figure 12.9(a) when the power of LO
is 0 dBm. Note that the on-chip generation of 0 dBm LO power by 65nm
CMOS has be en recently demonstrated in [176]. Compared to the subharmonic
mixer design in [234], the conve rsion gain is improved by more than 10 dB.
Also, the proposed mixer has a wide operation frequency range with a gain of
−19 ∼ −22dB from 220 GHz to 300 GHz. Moreover, a good input matching
and LO-RF isolation is a chieved with S11, S22 and S12 smaller than -10 dB
in 220 ∼ 300 GHz. Note that the conversion gain of the proposed mixer is also
determined by the available LO power at mixer input. As shown in Figure
12.9(a), the conversion ga in will drop to -37 dB when LO power is reduced to
-20 dBm.
12.3.3 Power Gain Amplifier
There are two major objectives in the PGA design in a THz heterodyne re-
ceiver. Fir st, sufficient gain must be provided to the targeted IF frequency
with a low noise figur e. Compared to the common-s ource amplifica tion topol-
ogy, casco de is more preferred with higher gain and stability. Second, a na r-
row frequency response is required to increase the selectivity of the receiver
270 Design of CMOS Millimeter-Wave and Terahertz Integrated Circuits
(a)
(b)
Figure 12.9: Simulation resu lts of propo sed mixer. (a) S-parameters
and conversion gain when s weeping RF and LO frequencies with
F
LO
= F
RF
+ 3GHz; (b) conversion gain at different LO power level
when sweeping RF frequency with F
IF
= F
RF
− 280GHz.
for the purpose of THz imaging. Generally, a narrow frequency response ca n
be achieved by a resona tor tank with a high quality factor, which is mainly
determined by the inductor for on-chip implementation. However, since the
inductor size is inversely proportio nal to the resonating frequency for a given
Q factor, it will gene rate large chip area overhead when the resonant frequency
is too small. As such, an optimized resonant frequency of 3GHz is selected in
the PGA design.
CMOS THz Imaging 271
Figure 12.10: Schematic of the three-stage power gain amplifier and
the output buffer.
Figure 12.10 shows the schematic of the proposed PGA, which is imple-
mented by three stages of cascode amplifiers followed by a common-source
output buffer. In each cascode stage, both transistors are biased in the sat-
uration region (0.6-V V
G
for M3, M5 and M7, V
G
is connected to VDD for
M4, M6 and M8.) The r esonator is implemented by a 410-fF metal-insulator-
metal capacitor and a 3.5nH spiral inductor. A common-source output buffer
is used to drive a 50-Ohm output impedance for the purpose of measurement.
The post-layout s imulation is also performed on the proposed PGA desig n.
As shown in Figure 12.11, a maximum gain of 33dB is obtained at a cen-
ter frequency of 3 GHz, and the 3-dB bandwidth is 150 MHz. Moreover, the
proposed PGA has a noise figure lower than 4 dB from 2.5 GHz to 3.5 GHz.
12.3.4 Wide-Band Imaging Results
The proposed wide-band CMOS imager is fabricated in Global Foundries (GF)
65-nm CMOS RF process after integrating the heterodyne receiver with the
circular polarized substrate integrated waveguide (SIW) antenna introduced
in Sec. 8 .3. The die microgra ph is shown in Figure 12.12 with a chip a rea of
0.99 mm
2
. The fabricated receiver chip is firstly measured alone followe d by
the applications in the THz imaging system.
12.3.4.1 Receiver Measurements
The receiver operates under 0.8-V powe r supply with overa ll p ower consump-
tion of 6.6 mW. As shown in Figure 12.13, the receiver chip is firstly measured
on a probe station (CASCADE Microtech Elite-300). An LO-signal (VDI) is
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